Stan Bauwens, Daan Vanden Meersschaut

powered by energy harvestingCompact transceiver for personal communication

Academiejaar 2012-2013Faculteit Ingenieurswetenschappen en ArchitectuurVoorzitter: prof. dr. ir. Daniël De ZutterVakgroep Informatietechnologie

Master in de ingenieurswetenschappen: elektrotechniekMasterproef ingediend tot het behalen van de academische graad van

Begeleider: Ramses PiercoPromotoren: prof. dr. ir. Johan Bauwelinck, prof. dr. ir. Jan Vandewege

Permission for use of content

“The authors give permission to make this master dissertation available for consulta-

tion and to copy parts of this master dissertation for personal use.

In the case of any other use, the limitations of the copyright have to be respected,

in particular with regard to the obligation to state expressly the source when quoting

results from this master dissertation.”

Stan Bauwens and Daan Van den Meersschaut, June 2013

iii

Toelating tot bruikleen

“De auteurs geven de toelating deze masterproef voor consultatie beschikbaar te stellen

en delen van de masterproef te kopieren voor persoonlijk gebruik.

Elk ander gebruik valt onder de beperkingen van het auteursrecht, in het bijzonder

met betrekking tot de verplichting de bron uitdrukkelijk te vermelden bij het aanhalen

van resultaten uit deze masterproef.”

Stan Bauwens en Daan Van den Meersschaut, juni 2013

iv

Stan Bauwens, Daan Vanden Meersschaut

powered by energy harvestingCompact transceiver for personal communication

Academiejaar 2012-2013Faculteit Ingenieurswetenschappen en ArchitectuurVoorzitter: prof. dr. ir. Daniël De ZutterVakgroep Informatietechnologie

Master in de ingenieurswetenschappen: elektrotechniekMasterproef ingediend tot het behalen van de academische graad van

Begeleider: Ramses PiercoPromotoren: prof. dr. ir. Johan Bauwelinck, prof. dr. ir. Jan Vandewege

Preface

This thesis is the completion of our five year long education in electrical engineering.It was an ideal chance to design a practical solution for a system for which there is alot of future. We would like to thank prof. dr. ir. Johan Bauwelinck and prof. dr. ir.Jan Vandewege to have given us this opportunity.

Without the assistance of our supervisor ir. Ramses Pierco, this thesis would not havebeen this successful and educational. We are very grateful for his support and guidanceand we very much appreciated his feedback on our report.

A special thanks goes to ing. Jan Gillis and ir. Li Xiao. Jan Gillis for the numerouscreated printed circuit boards and shared experience concerning soldering. Li Xiao forhis help during the difficult periods of debugging code.We would also like to thank the rest of the INTEC DESIGN research group to createan open environment where help was only a small step away.

Family and friends also gave us moral support and stimulation to continue the goodwork, in particular Delphine Vanvooren. For this we would like to thank them.

We thank each other. It was very useful to do this thesis together. A lot more couldbe achieved than when working alone.

Last but not least, we would like to honour our colleagues at the other side of theroom, Sander Lybeert and Marijn Verbeke. They were always ready to answer everylittle question and made the ’thesiskot’ a place where you could work in a nice am-biance. Thank you for this wonderful year, gentlemen.

Stan Bauwens and Daan Van den Meersschaut, June 2013

vi

Compact Transceiver for Personal Communication

Powered by Energy Harvesting

by

Stan Bauwens & Daan Van den Meersschaut

Master’s Thesis submitted to obtain the academical degree ofMaster of Science in Electrical Engineering

Academic year 2012-2013

Promotors: prof. dr. ir. Johan Bauwelinck, prof. dr. ir. Jan VandewegeSupervisor: ir. Ramses Pierco

Faculty of Engineering and ArchitectureGhent University

Department of Information TechnologyChairman: prof. dr. ir. Daniel De Zutter

Summary

In this master dissertation a fully autonomous wireless sensor module is designed. Au-tonomy is guaranteed using energy harvesting to power the sensor module. In the firstchapter a possible application is discussed. The following chapter handles the conceptof energy harvesting. The four covered energy sources are solar, thermal, vibration andRF. The conclusion of these sections is that solar and thermal energy harvesting gene-rate the most power and will serve as power supply. In the third chapter the storageof energy using supercapacitors and thin-film batteries is described. In the followingchapter, the implementation of the transceiver of the sensor module is discussed whichconsists of a microcontroller, an antenna chip and a sensor. Furthermore a base stationwith the same components is designed which communicates with the computer. Thereceived data is then visualised. The last chapter combines all the previous chaptersto create an operational system.

Keywords

Energy Harvesting, Wireless, Sensor, Thin-film Battery, Supercapacitor

Compact transceiver for personal communicationpowered by energy harvesting

Stan Bauwens & Daan Van den Meersschaut

Supervisor(s): prof. dr. ir. Johan Bauwelinck, prof. dr. ir. Jan Vandewege and ir. Ramses Pierco

Abstract— In this article, a practical solution for an autonomous wire-less sensor network is explained. It is powered using Energy Harvesting toguarantee its autonomy.

Keywords— Energy Harvesting, Wireless Transceiver, Sensor, Thin-filmBattery, Supercapacitor, Low Power

I. INTRODUCTION

THE problem of monitoring elderly people has been a hottopic for the past few years. For these people autonomy is

very important for their mental well-being which leads to a needfor an autonomous monitoring system. A possible solution forthis problem is discussed in this article, an autonomous wire-less sensor module. The sensor module is powered using en-ergy harvesting (EH). Due to the limited available power, all thecomponents should be consuming as little energy as possible.Therefore the sensor module only sends data periodically. Be-tween two transmissions the sensor module goes to a low powermode (LPM) to safe energy.

II. ENERGY HARVESTING

Energy harvesting is vital for the autonomous operation ofsensor modules. Battery replacement is not necessary when en-ergy is continuously gathered from different external sources.Solar energy, thermal energy, vibrational energy and energy con-tained in RF waves are the different energy types discussed.

A. Solar

A solar energy harvesting circuit was designed, using theLTC3105 DC/DC converter from Linear Technology. Solar en-ergy gathered by the SLMD121H10 solar module from IXYS isstored by the LTC3105 on a 3 F supercapacitor, the supercapac-itor is charged to 3.3 V. The efficiency of the LTC3105 stronglydepends on the output capacitor voltage. To improve efficiencythe supercapacitor should not be discharged below 2 V. Since aconstant supply voltage of 3.3 V is needed and the supercapaci-tor voltage depends on the stored energy, a step-up converter isneeded. The TPS61221 of Texas Instruments is used. With 2 Vas the minimal supercapacitor voltage, a total charge/dischargeefficiency of 79.9 % is reached. At an illuminance of 65 klux, anaverage power of 77 mW (7mW/cm2) is stored on the superca-pacitor. Figure 1 shows the average storage power as a functionof the capacitor start voltage of the charge cycle (Vcap,start).

B. Thermal

If the sensor module is a personal device, body heat is alsoa possible source of energy. For indoor applications, solar en-ergy produces little energy. The LTC3109 from Linear Tech-nology is a low input voltage DC/DC converter and is used to

Fig. 1. Average stored power as a function of the supercapacitor start voltage ofthe charge cycle (Vcap,start) at an illuminance of 65 klux

Fig. 2. The output power as a function of load impedance for the PolarTEC PT4at a temperature difference of 20C

convert the low-voltage power generated by a peltier element.The peltier element converts thermal energy into electric energybut the voltages generated can be very low, especially when us-ing body heat as an energy source. At a 20C temperature dif-ference, the maximum available power generated by the peltierelement is 3.6 mW (cf. figure 2). Under optimal conditionsthe LTC3109 reaches an efficiency of 28.3 %. At this maximumpower point a power of 480 µW (47µW/cm2) is available at theload when a realistic temperature difference of 10C is applied.

C. Vibration

Due to the portability of the sensor module, vibrational energyis also a possible power source. A piezoelectric element is usedto generate electric power. The resulting AC voltages have tobe rectified and converted to a proper usable DC voltage. TheLTC3588-1 chip is used to do this conversion.

Fig. 3. Spectrum below 1 GHz, Ghent

The power which is captured from this energy source is low (4µW , cf. [1], [2]) in comparison to solar energy and thermalenergy, so for the end system, this type of energy is not used.

D. RF

Capturing radio waves is ever more interesting due to the in-crease of wireless transmissions. Radio waves above 1 GHz ingeneral carry less peak power than radio waves below 1 GHz.The spectrum from 10 MHz to 990 MHz has multiple interest-ing frequency bands (cf. figure 3). The bands have an averagetotal channel power of approximately -40 dBm (except FM -33.5dBm).

A rectangular patch antenna is used to capture the RF signals.The dimensions are proportional to the wavelength. This makeslow frequencies hard to capture with a small antenna. Anothersubstrate with a higher εr can be used to decrease the dimen-sions. However this will lead to a worse transmission coefficientfrom the air to the substrate.SO3010 is a high frequency PCB material with an εr of 11.2.The material is characterised using simulations and measure-ment to have an εr of 10.7. The power transmission from the airto this substrate is a factor 2 smaller than FR4. This makes, thissubstrate a viable solution if the power is twice as high at lowerfrequencies.A thicker substrate will emit more radiation than a thin, due tomore fringe fields at the side of the antenna.It is possible (under right circumstances) to use RF energyto power a device, but for the portable device with a certainsendrate (once per 10 s/1 min) the power (0.1 - 1 µW cf. [1],[2]) is too small.

III. STORAGE

The generated power is only partially used by the transceiver.The remaining energy has to be stored for periods during whichno energy is available from energy harvesting. The energy har-vested by solar energy is stored in a 3 F supercapacitor (cf. fig-ure 4). Because the voltage over this capacitor isn’t constant, aboost converter is needed to convert the voltage to a constant 3.3V supply. The used converter is the TPS61221 from Texas In-struments. The energy harvested from thermal energy is storedby the LTC3109 on a 0.33 F storage capacitor. When the thermal

source becomes unavailable, energy from this storage capacitoris used by the LTC3109 to power the output by an internal buckconverter. No modules have been made to harvest energy fromvibrational or RF energy.

Because two or more energy sources can be used at the sametime, a power combination circuit is implemented. To this endtwo CBC3150 EnerChips from Cymbet are used. These com-ponents consist of a thin-film battery and an internal controlcircuit and charge pump. The batteries of the two EnerChipsare made common and each EnerChip can charge the commonbattery with the energy from its input. The outputs of the En-erChips are connected via two diodes. To handle large currentpulses sinked by the digital part of the system, boost capacitorswith a total value are 1.41 mF are added (cf. figure 4).

IV. TRANSCEIVER

The generated and stored energy is used to power atransceiver which consists of three mainly digital parts: a micro-controller, an antenna chip (with antenna) and a sensor. Thesethree components plus the storage and EH part, form the sensormodule. There is also another module, the base station, whichconsists of the same three components but is powered using theUSB-port of the computer. This is shown in figure 5.

A. Microcontroller

The microcontroller (MSP430G2553) communicates with thesensor and antenna chip using SPI (Serial Peripheral Interface).The microcontroller is the master and selects a slave to commu-nicate with by pulling the correct chip select pin low. The mastersets the registers of the slaves in order to let them function cor-rectly.The microcontroller also determines the timetable of the sensormodule. At a regular interval, the microcontroller will awakenitself and the antenna chip. It will read and process data (sensordata and circuit voltage levels), and afterwards sends this data tothe antenna chip who will transmit it wirelessly.

B. Sensor

In the context of the monitoring application, the sensor is anaccelerometer. The sensor has to be able to detect an event ofpeople falling. If this happens an interrupt is sent to the micro-controller. Note that the sensor has to be active at all times toprevent a missed event.

C. Transmission protocol

The carrier frequency of the antenna chip (CC110L) is 433MHz which lies in the ISM band. The sensor module will alsoawake and transmit if the microcontroller gets an interrupt froma button or the accelerometer. After every transmission, the sen-sor module listens for a reply for 10 ms. Afterwards, the micro-controller processes the received data (or does nothing if nothingwas received) and puts the sensor module back to sleep (LPM).

Solar cell Regulator

Super-Capacitor

Step-up converter

Thin-film batteries

Boost-Capacitor

Microcontroller

Peltier element Regulator

Super-Capacitor

Sensor

Antenna chip

Thermal board

Solar board

Sensor module board

Fig. 4. Schematic of the complete sensor module

MicrocontrollerEnergy Harvesting

Sensor Module

SPI bus

Antenna Chip

Sensor __CS

__CS

Storage

Base Station

Antenna Chip

__CS

__CS

MicrocontrollerSPI busUART

Fig. 5. Schematic of the complete system

At the receiving end, the base station receives a message froma sensor module and sends back a short reply, possible contain-ing settings for the corresponding sensor module. The receiveddata from the sensor modules is sent from the BS to the PC us-ing UART (Universal asynchronous receiver/transmitter).

D. GUI

The data is then visualised by a Graphical User Interface(GUI) in MATLAB. This data contains the SensorID from thecorresponding sensor module, the voltage level of its battery andtwo bits which indicate a fall or button interrupt.

V. CONCLUSION

A fully functional system is created with autonomous sensormodules. The data is correctly received and displayed. Thereare however improvements needed to bring this product on themarket. For example: further miniaturisation, improvements ofthe communication protocol, optimising the settings of compo-nents, etc.

REFERENCES

[1] Murugavel Raju, Mark Grazier White Paper, Energy Harvesting,http://www.ti.com/lit/wp/slyy018a/slyy018a.pdf

[2] Faruk Yildiz Potential Ambient Energy-Harvesting Sources and Techniques,http://scholar.lib.vt.edu/ejournals/JOTS/v35/v35n1/pdf/yildiz.pdf

[3] Stan Bauwens, Daan Van den Meersschaut Compact transceiver for per-sonal communication powered by energy harvesting

Een compacte transceiver voor persoonlijkecommunicatie gevoed door energy harvesting

Stan Bauwens & Daan Van den Meersschaut

Supervisor(s): prof. dr. ir. Johan Bauwelinck, prof. dr. ir. Jan Vandewege en ir. Ramses Pierco

Abstract—In dit artikel, wordt een praktische oplossing voor autonomedraadloze sensornetwerken uitgelegd. Om de autonomie van de individuelemodules te garanderen, worden ze gevoed met behulp van energy harvest-ing.

Keywords— Energy Harvesting, Draadloze Zender, Sensor, Dunne-filmBatterij, Supercapaciteit, Laag Vermogen

I. INTRODUCTIE

HET probleem van het toezicht op ouderen is een hot topicin de afgelopen jaren. Autonomie is erg belangrijk voor

hun geestelijk welzijn, wat leidt tot een behoefte voor een au-tonoom observatiesysteem. Een mogelijke oplossing voor ditprobleem wordt besproken in dit artikel, een autonome draad-loze sensormodule. De sensormodule wordt gevoed met behulpvan energy harvesting (EH). Vanwege het beperkt beschikbaarvermogen moeten alle onderdelen zo weinig mogelijk energieverbruiken. Daarom verstuurt de sensormodule data periodiek.Tussen twee transmissies gaat de sensormodule in een laagver-mogenmodus om energie te besparen.

II. ENERGY HARVESTING

Energy harvesting is van vitaal belang voor de autonomewerking van de sensormodules. Het vervangen van batterijenis niet nodig wanneer energie continu wordt verzameld uit ver-schillende externe bronnen. Zonne-energie, thermische energie,vibratie-energie en energie uit radiogolven zijn de verschillendeenergytypes die besproken worden.

A. Zonne-energie

Een circuit voor het opslaan van zonne-energie is ontworpenmet behulp van de LTC3105 DC/DC-omzettter van Linear Tech-nology. Zonne-energie verzameld door de SLMD121H10 zon-nemodule van IXYS wordt opgeslagen door de LTC3105 op eensupercondensator van 3 F, de supercondensator wordt opgeladentot 3.3 V. De efficientie van de LTC3105 is sterk afhankelijk vande condensatorspanning. Om de efficientie te verbeteren magde supercondensator niet worden afgeladen tot onder de 2 V.Aangezien een constante voedingsspanning van 3.3 V nodig isen de supercondensatorspanning afhankelijk is van de opgesla-gen energie, is een step-up omzetter nodig. Hiertoe wordt deTPS61221 van Texas Instruments gebruikt.Met 2 V als de minimale supercondensatorspanning, wordt eentotale laad/ontlaad-efficientie van 79.9 % is bereikt. Bij een il-luminantie van 65 klux, wordt een gemiddeld vermogen van 77mW (7 mW/cm2) opgeslagen op de supercondensator. Figuur1 toont het gemiddeld opslagvermogen als functie van de con-densator startspanning van de laadcyclus.

Fig. 1. Gemiddeld opgeslagen vermogen als functie van de supercondensatorstartspanning van de oplaadcyclus (Vcap,start) bij een illuminantie van 65klux

Fig. 2. PolarTEC PT4 uitgangsvermogen als functie van lastimpedantie bij eentemperatuurverschil van 20C

B. Thermische energie

Als de sensormodule kort op het lichaam gehouden kan wor-den, is lichaamswarmte een mogelijke bron van energie. Voorbinnenshuistoepassingen is er niet genoeg energie beschikbaaruit zonne-energie. De LTC3109 van Linear Technology is eenlage ingangsspanning DC/DC-omzetter en wordt gebruikt omde zeer lage spanningen opgewekt door een peltier element omte zetten naar een vaste 3.3 V. Het peltier element zet thermis-che energie om in elektrische energie maar de spanningen kun-nen zeer laag zijn, zeker als lichaamswarmte als energiebronwordt gebruikt. Bij een temperatuurverschil van 20C, bedraagthet maximaal vermogen gegenereerd door het peltierelement 3.6mW (cf. figuur 2). Onder optimale omstandigheden bereikt deLTC3109 een rendement van 28.3 %. Een maximaal vermogenvan 480 µW (47 µW/cm2) is bij een temperatuurverschil van

Fig. 3. Spectrum onder 1 GHz, Gent

10C beschikbaar aan de last.

C. Vibratie-energie

Door de draagbaarheid van de sensormodule is vibratie-energie ook een mogelijke energiebron. Een piezo-elektrischelement wordt gebruikt om elektrische energie te genereren.De resulterende wisselspanning moet gelijkgericht en omgezetworden in een goed bruikbare constante voedingsspanning. DeLTC3588-1 chip wordt gebruikt om deze conversie uit te voeren.

Het ontvangen vermogen van deze energiebron is laag (4 µW[1], [2]) in vergelijking met zonne-energie en thermische en-ergie. Om deze reden wordt piezo-elektrische energie niet ge-bruikt in het systeem.

D. RF

Energie halen uit radiogolven wordt steeds interessanter, ditis te wijten aan de toename van radiogolven in de lucht. Ra-diogolven met frequenties boven 1 GHz bevatten in het alge-meen minder vermogen dan radiogolven met frequenties onderde 1 GHz. Het lokale spectrum [10 MHz, 990 MHz] (in Gent)heeft meerdere interessante frequentiebanden (zie figuur 3). Debanden hebben elk een gemiddeld totaal kanaalvermogen vanongeveer -40 dBm (uitgezonderd FM met een kanaalvermogenvan -33.5 dBm).Een rechthoekige patch-antenne wordt gebruikt om RF-signalente ontvangen. De afmetingen van de antenne zijn evenredig metde golflengte. Dit maakt dat lage frequenties moeilijker te ont-vangen zijn met een kleine antenne. Een ander substraat met eenhogere εr kan worden gebruikt om de afmetingen te verkleinen.Dit zal leiden tot een slechtere transmissiecoefficient tussen delucht en het substraat.SO3010 is een PCB-materiaal voor hoge frequenties met een εrvan 11.2. Aan de hand van simulaties en metingen bleek dat hetsubstraat een εr heeft van 10.7. De vermogenoverdracht van delucht naar het substraat is een factor 2 kleiner dan bij het FR4substraat. Dit maakt het substraat pas een goede oplossing alshet vermogen bij de nu bereikbare lagere frequenties twee keerzo hoog is als bij hogere frequenties.Een dikker substraat zendt meer straling uit dan een dun sub-straat doordat er meer franjevelden zijn.Het is mogelijk (onder de juiste omstandigheden) om RF-

energie te gebruiken voor het voeden van sommige systemen.Voor het draagbare systeem hier besproken met een sendratevan 1 transmissie per 10 s/1 min, is het vermogen (0.1-1 µW[1], [2]) te klein.

III. ENERGIE-OPSLAG

De gewonnen energie wordt slechts gedeeltelijk door desensormodule gebruikt. De resterende energie moet wordenopgeslagen voor momenten wanneer er geen energie meer kanopgewekt worden uit de gebruikte bronnen. De energie gewon-nen uit zonne-energie wordt opgeslagen in een superconden-sator van 3 F. Omdat de spanning over deze condensator nietconstant is, wordt een boost-omzetter gebruikt om de span-ning om te zetten in een constante 3.3 V voedingsspanning.De gebruikte omzetter is de TPS61221 van Texas Instruments.De gewonnen thermische energie wordt opgeslagen door deLTC3109 op een 0.33 F opslagcondensator. Wanneer de ther-mische bron niet meer beschikbaar is, wordt de energie uit dezeopslagcondensator gebruikt door de LTC3109 om de last aan deoutput te voeden. Dit gebeurt aan de hand van een interne buck-omzetter.

Omdat 2 of meer energiebronnen tegelijkertijd moeten kun-nen worden gebruikt, wordt een vermogencombinatie circuitontworpen. Hiervoor worden 2 CBC3150 EnerChips van Cym-bet gebruikt. Deze componenten bestaan uit een dunne-film bat-terij en een intern regelcircuit en ladingspomp. De batterijen vande 2 EnerChips zijn gemeenschappelijk gemaakt en elke Ener-Chip kan de gemeenschappelijke batterij opladen met de energiebeschikbaar aan zijn ingang. De uitgangen van de EnerChipszijn via 2 diodes verbonden met elkaar. Om grote stroompulsenvan de microcontroller aan te kunnen, zijn boostcondensator meteen totale waarde van 1.41 mF toegevoegd.

IV. TRANSCEIVER

De gegenereerde en opgeslagen energie wordt gebruikt voorhet aandrijven van een transceiver die bestaat uit drie hoofdza-kelijk digitale onderdelen: een microcontroller, een antennechip(met antenne) en een sensor. Deze drie componenten, de op-slagelementen en het EH deel vormen de sensormodule. Er isook een andere module, het basisstation, die bestaat uit dezelfdedrie componenten, maar gevoed wordt via de USB-poort van decomputer. Dit wordt weergegeven in figuur 5.

A. Microcontroller

De microcontroller (MSP430G2553) communiceert met desensor en antennechip met behulp van SPI (Serial Peripheral In-terface). De microcontroller is de master en selecteert een slavemet behulp van de juiste chip select pin. De master stelt de reg-isters van de slaves in om ze correct functionerend te houden.De microcontroller bepaalt ook het tijdschema van de sensor-module. Met regelmatige tussenpozen zal de microcontrollerzichzelf en de antennechip doen ontwaken. De microcon-troller zal gegevens lezen en verwerken, en daarna stuurt het degegevens naar de antennechip die de data draadloos verstuurt.

Solar cell Regulator

Super-Capacitor

Step-up converter

Thin-film batteries

Boost-Capacitor

Microcontroller

Peltier element Regulator

Super-Capacitor

Sensor

Antenna chip

Thermal board

Solar board

Sensor module board

Fig. 4. Schema van de volledige sensormodule

B. Sensor

In het kader van de toezichttoepassing is de sensor een ac-celerometer. De sensor moet in staat zijn om een gebeurtenisvan vallende mensen te detecteren. Als dit gebeurt, wordt eeninterrupt gezonden naar de microcontroller. Merk op dat de sen-sor de hele tijd actief dient te zijn om een gemiste gebeurtenis tevoorkomen.

C. Transmissie protocol

De draaggolffrequentie van de antennechip (CC110L) is 433MHz, deze frequentie ligt in de ISM band. De sensormodule zalook ontwaken en data verzenden als de microcontroller een in-terrupt krijgt van een drukknop of van de versnellingsmeter. Naelke verzending, wacht de sensormodule 10 ms op een antwo-ord. Daarna verwerkt de microcontroller de ontvangen data (ofdoet niets als er niets is ontvangen) waarna de sensormoduleterug naar laag vermogen modus gaat.

Aan de andere kant, ontvangt het basisstation een bericht enstuurt het een kort antwoord met mogelijk een instellingswijzig-ing voor de bijhorende sensormodule. De ontvangen gegevensvan de sensormodules worden verzonden van het basisstationnaar de PC met behulp van UART (Universal asynchronous re-ceiver/transmitter).

D. GUI

De gegevens worden vervolgens gevisualiseerd met een GUI(Graphical User Interface) in MATLAB. De gegevens bevatten

de SensorID van de overeenkomstige sensormodule, het span-ningsniveau van de batterij en bits die een val of knop interruptaanduiden.

V. CONCLUSIE

Een volledig functionerend systeem met autonome sensor-modules is ontworpen. De gegevens worden correct ontvangenen weergegeven. Er zijn echter verbeteringen nodig om dit prod-uct op de markt te brengen. Bijvoorbeeld: verdere miniaturis-ering, verbetering van het communicatieprotocol, optimaliserenvan de instellingen van de componenten, etc.

REFERENCES

[1] Murugavel Raju, Mark Grazier White Paper, Energy Harvesting,http://www.ti.com/lit/wp/slyy018a/slyy018a.pdf

[2] Faruk Yildiz Potential Ambient Energy-Harvesting Sources and Techniques,http://scholar.lib.vt.edu/ejournals/JOTS/v35/v35n1/pdf/yildiz.pdf

[3] Stan Bauwens, Daan Van den Meersschaut Compact transceiver for per-sonal communication powered by energy harvesting

MicrocontrollerEnergy Harvesting

Sensor Module

SPI bus

Antenna Chip

Sensor __CS

__CS

Storage

Base Station

Antenna Chip

__CS

__CS

MicrocontrollerSPI busUART

Fig. 5. Schema van het volledige systeem

Contents

Permission for use of content iii

Toelating tot bruikleen iv

Preface vi

Summary vii

Extended abstract vii

Extended abstract Dutch xi

Contents xiv

List of Figures xvii

List of Tables xx

Glossary xx

1 Introduction 1

2 Application 2

3 Energy Harvesting 4

3.1 Solar . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

3.1.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

3.1.2 Solar module . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

3.1.3 Regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

3.1.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

3.2 Thermal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3.2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3.2.2 Peltier element . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

3.2.3 Regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

3.2.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

xiv

3.3 Vibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.3.2 Piezoelectric element . . . . . . . . . . . . . . . . . . . . . . . . 18

3.3.3 Full wave rectifier and buck converter . . . . . . . . . . . . . . . 19

3.3.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.4 RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.4.2 Possibilities . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.4.3 Simulations and measurements . . . . . . . . . . . . . . . . . . 27

3.4.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

4 Storage 33

4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

4.2 Storage methods . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

4.2.1 Conventional rechargeable batteries . . . . . . . . . . . . . . . . 33

4.2.2 Supercapacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

4.2.3 Thin-film batteries . . . . . . . . . . . . . . . . . . . . . . . . . 37

4.2.4 Choice of the storage component . . . . . . . . . . . . . . . . . 38

4.3 Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

4.3.1 Thin-film battery . . . . . . . . . . . . . . . . . . . . . . . . . . 39

4.3.2 Supercapacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . 42

4.3.3 Pulse current applications . . . . . . . . . . . . . . . . . . . . . 49

4.3.4 Storage circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

5 Transceiver 57

5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

5.2 Functional overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58

5.3 Sensor module . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58

5.3.1 Microcontroller . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

5.3.2 Antenna chip . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

5.3.3 Sensor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

5.4 Implementation sensor module . . . . . . . . . . . . . . . . . . . . . . . 65

5.4.1 Connections and communication . . . . . . . . . . . . . . . . . . 65

5.4.2 Settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

5.4.3 Coding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

5.4.4 Coding and testing issues . . . . . . . . . . . . . . . . . . . . . 74

5.4.5 Board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

5.4.6 Power performance . . . . . . . . . . . . . . . . . . . . . . . . . 75

5.5 Base station . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

5.5.1 Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

5.6 Implementation BS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

xv

5.6.1 Connections and communication . . . . . . . . . . . . . . . . . . 78

5.6.2 Settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

5.6.3 Coding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79

5.7 Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

5.7.1 Communication . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

5.7.2 Power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

5.8 Matlab . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

5.9 Improvements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84

6 Total system 90

6.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

6.2 Boards . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90

6.3 Performance and measurements . . . . . . . . . . . . . . . . . . . . . . 91

7 Conclusion 94

A Figures and Layout PCBs 95

A.1 Boards of the complete sensor module . . . . . . . . . . . . . . . . . . . 95

A.1.1 Solar board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95

A.1.2 Thermal board . . . . . . . . . . . . . . . . . . . . . . . . . . . 96

A.1.3 Sensor module board . . . . . . . . . . . . . . . . . . . . . . . . 97

A.2 Test boards . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98

Bibliography 102

xvi

LIST OF FIGURES xvii

List of Figures

3.1 Output power density of the CPC1824 and SLMD121H10 under different

loads at an illuminance of 12 klux . . . . . . . . . . . . . . . . . . . . . 5

3.2 Output power of the SLMD121H10 as a function of load resistance at

different light intensities . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.3 Output resistance of the SLMD121H10 solar module as a function of

illuminance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.4 Schematic of the solar energy harvesting circuit . . . . . . . . . . . . . 8

3.5 LTC3105 power efficiency under different output voltages at Vin=2.6 V

(illuminance = 65 klux) . . . . . . . . . . . . . . . . . . . . . . . . . . 9

3.6 Output and input power of the LTC3105 at an illuminance of 65 klux . 10

3.7 Total energy efficiency of a charge cycle as a function of capacitor start

voltage (Vcap,start) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

3.8 Average stored power as a function of the supercapacitor start voltage

of the charge cycle (Vcap,start) at an illuminance of 65 klux . . . . . . . . 12

3.9 Output power of the PolarTECTM PT4 as a function of the temperature

difference at different loads (1.2 kΩ and 12 kΩ) . . . . . . . . . . . . . 14

3.10 The Output power as a function of load impedance for the PolarTECTM

PT4 at a temperature difference of 20C . . . . . . . . . . . . . . . . . 14

3.11 Schematic of the thermal energy harvesting circuit . . . . . . . . . . . . 17

3.12 LTC3109 power efficiency under different input voltages at a load of 12kΩ 18

3.13 Piezoelectric element: V22BL . . . . . . . . . . . . . . . . . . . . . . . 20

3.14 Schematic of the piezoelectric energy harvesting circuit . . . . . . . . . 20

3.15 Spectrum below 1 GHz, Ghent . . . . . . . . . . . . . . . . . . . . . . . 23

3.16 Spectrum FM-Radio, Span 35 MHz . . . . . . . . . . . . . . . . . . . . 24

3.17 Spectrum T-DAB, Span 35 MHz . . . . . . . . . . . . . . . . . . . . . . 24

3.18 Spectrum DVB-T at 482 MHz, Span 35 MHz . . . . . . . . . . . . . . . 24

3.19 Spectrum DVB-T at 650 MHz, Span 70 MHz . . . . . . . . . . . . . . . 24

3.20 Spectrum GSM, Span 70 MHz . . . . . . . . . . . . . . . . . . . . . . . 25

3.21 Patch antenna, fringe fields . . . . . . . . . . . . . . . . . . . . . . . . . 27

3.22 FR4 patch: S(6,6) simulation, S(5,5) measurement . . . . . . . . . . . . 27

3.23 Simulation RO3010 patches: S(2,2) 1.28 mm, S(4,4) 0.64 . . . . . . . . 28

3.24 SO3010 patch (1.28 mm): S(4,4) simulation, S(3,3) measurement . . . . 29

3.25 SO3010 patch (0.64 mm): S(2,2) simulation, S(1,1) measurement . . . . 29

3.26 Measurement SO3010 patches: S(3,3) 1.28 mm, S(1,1) 0.64 mm . . . . 29

3.27 Simulation optimised SO3010 patches: S(1,1) 1.28 mm, S(2,2) 0.64 mm 30

4.1 Ragone chart [28] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

4.2 Typical EnerChip discharge characteristic as copied from datasheet CBC3150

[29] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

4.3 Battery voltage and Vout of two battery-connected CBC3150 . . . . . . 43

4.4 Input charge current as a function of the battery voltage . . . . . . . . 43

4.5 Supercapacitors as storage element . . . . . . . . . . . . . . . . . . . . 46

4.6 TPS61221 power efficiency as a function of input voltage, at an output

current of 95 µA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

4.7 Thin-film battery, boost capacitor and load during pulses . . . . . . . . 49

4.8 Charging of the boost capacitor with a single thin-film battery . . . . . 50

4.9 Cell resistance as a function of state of charge (Cymbet application note

[34]) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

4.10 EnerChip circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

4.11 Input current as a function of time . . . . . . . . . . . . . . . . . . . . 56

4.12 Output power when discharging the thin-film batteries with a 67.5kΩ load 56

5.1 Schematic of the complete system . . . . . . . . . . . . . . . . . . . . . 57

5.2 Launchpad Development Tool with MSP430G2553 . . . . . . . . . . . . 61

5.3 CC110L AIR Module BoosterPack . . . . . . . . . . . . . . . . . . . . 63

5.4 Launchpad with Boosterpack plugged in . . . . . . . . . . . . . . . . . 63

5.5 ADXL362 Schematic overview . . . . . . . . . . . . . . . . . . . . . . . 65

5.6 ADXL362 Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

5.7 Configuration pins of the µC, fixed SPI and analog inputs . . . . . . . 67

5.8 Active Mode Current vs DCO Frequency . . . . . . . . . . . . . . . . . 68

5.9 Schematic overview code for the sensor module . . . . . . . . . . . . . . 72

5.10 Simple board sensor module with accelerometer . . . . . . . . . . . . . 75

5.11 Simple board sensor module without accelerometer . . . . . . . . . . . 75

5.12 Current profile sensor module, BS in range, 0 dBm . . . . . . . . . . . 77

5.13 Current profile sensor module, no BS in range, 0 dBm . . . . . . . . . . 77

5.14 Schematic overview of the code for the base station . . . . . . . . . . . 81

5.15 Part 1: SPI and UART communication of SM (D0-D3) and BS (D8-D13) 83

5.16 Part 2: SPI and UART communication of SM (D0-D3) and BS (D8-D13) 83

5.17 GUI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85

5.18 Protocol timeline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

6.1 Complete sensor module (top view) . . . . . . . . . . . . . . . . . . . . 91

6.2 Complete sensor module (side view) . . . . . . . . . . . . . . . . . . . . 91

6.3 Schematic overview of the sensor module and its supply modules . . . . 91

xviii

6.4 Visualisation in Matlab of 2 sensor modules in a network . . . . . . . . 93

A.1 Solar energy harvesting board, PCB layout top layer . . . . . . . . . . 95

A.2 Solar energy harvesting board, PCB layout bottom layer . . . . . . . . 95

A.3 Solar energy harvesting board . . . . . . . . . . . . . . . . . . . . . . . 96

A.4 Thermal energy harvesting board, PCB layout top layer . . . . . . . . . 96

A.5 Thermal energy harvesting board, PCB layout bottom layer . . . . . . 96

A.6 Thermal energy harvesting board . . . . . . . . . . . . . . . . . . . . . 97

A.7 Sensor module board, PCB layout top layer . . . . . . . . . . . . . . . 97

A.8 Sensor module board, PCB layout bottom layer . . . . . . . . . . . . . 98

A.9 Sensor module board . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98

A.10 Test board for solar energy harvesting (LTC3105) . . . . . . . . . . . . 99

A.11 Test board for thermal energy harvesting (LTC3109) . . . . . . . . . . 99

A.12 Test board for vibrational energy harvesting (LTC3588-1) . . . . . . . . 100

A.13 Test board for the thin-film batteries (CPC3150) . . . . . . . . . . . . 100

A.14 Active diode (LTC4413) . . . . . . . . . . . . . . . . . . . . . . . . . . 101

xix

LIST OF TABLES xx

List of Tables

3.1 Characterisation SO3010 substrate permittivity . . . . . . . . . . . . . 30

5.1 Properties of MSP430G2553 . . . . . . . . . . . . . . . . . . . . . . . . 61

5.2 Properties of CC110L . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63

5.3 Properties of ADXL362 . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

5.4 Allocation of the data lines . . . . . . . . . . . . . . . . . . . . . . . . . 82

5.5 Transmit power versus receive range at 433 MHz in an outdoor situation 85

Glossary

µC Microcontroller

ACLK Auxiliary clock

ADC Analogue to Digital Converter

ASK Amplitude-shift Keying

BS Base Station

BSL Bootstrap Loader

BW Bandwidth

CCS Code Composer Studio

CLK Clock

COM Serial Communication Port

CS Chip Select

DCO Digital Controlled Oscillator

DVB-T Digital Video Broadcast Terrestrial

ECC Electronic Communications Committee

EH Energy Harvesting

FCC Federal Communications Commission

FIFO First In First Out

GFSK Gaussian Frequency-shift Keying

GUI Graphical User Interface

ISM Industrial, Scientific, Medical

xxi

LPM Low Power Mode

MISO Master In Slave Out

MOSI Master Out Slave In

MSP Signal Microprocessor

PCB Printed Circuit Board

PMR Personal Mobile Radio

SM Sensor Module

SME Small or Medium Enterprise

SPI Serial Peripheral Interface

T-DAB Terrestrial-Digital Audio Broadcast

TI Texas Instruments

UART Universal asynchronous Receiver

and Transmitter

xxii

INTRODUCTION 1

Chapter 1

Introduction

The assignment for this thesis is to create a compact transceiver for personal com-

munication (a sensor module). This sensor module has to be powered using energy

harvesting. Due to this harvesting the sensor module becomes completely autono-

mous. Autonomy ensures a broad application area. A whole network of these modules

would enable fine-grained, long-term monitoring without the need of tens or hundreds

of battery-powered nodes to be deployed. A network with a large number of nodes,

makes battery replacement a cumbersome and time consuming task.

The main focus of this thesis is to make a practical (small) sensor network which has

(for now limited) communication possibilities and does not need external adjustments

to function correctly after the system has started up. Monitoring the operations of the

network should be possible.

Monitoring environmental properties is a possible application. Afterwards a central

system can use this information to control and monitor the environment or adapt itself

according to the changing properties. If these properties represent the state of certain

people carrying along the sensor modules, the modules become personal transceivers.

In the next chapter such an application is discussed, upon which some of the design

choices are based.

APPLICATION 2

Chapter 2

Application

For this thesis, an independent wireless sensor network is created which communicates

with a fixed base station. The main advantage of using a wireless system is the porta-

bility and autonomy of the sensor modules. This makes it ideal for monitoring systems.

A promising application of this is situated in the health care sector 1.

Monitoring elderly people is a hot topic due to the ageing of the population. There is

a growing need to monitor these people and to wirelessly tranfer alarm signals when

accidents occur. The current products for personal portable alarms and constant mo-

nitoring are mainly based on batteries which have to be replaced every now and then.

Also, for personal alarms, there is typical no guarantee or warning when going out of

range of the alarm detector. For these reasons, our system can perform better.

Our portable system does not need any external power sources and can send data

frequently if desired. Multiple monitoring functions are possible: skin temperature,

movement (fall sensor), sound, etc. Other human processes (heartbeat, certain blood

levels, ...) can also be monitored, however there is no commercially low power sensor

available for this.

The signal strength and absence of the regular burst of data can be used to do an out-

of-range-check and a life-check. This way, the chance of not detecting a malfunction

can be diminished.

To increase the operational range multiple base stations can be installed. This can

lower the power consumption of the sensor module since a smaller transmission power

is needed when the distance to the base station(s) is decreased. This can also benefit

localisation of the module, when we compare the different received powers in multiple

base stations.

1This application was based on an idea from Alphatronics, an SME based in Lokeren. They alreadydetermined and analysed the economic feasibility.

APPLICATION 3

The base stations are able to receive multiple signals and to distinguish between them.

This makes multi-user applications possible. This is very useful in rest homes where

monitoring certain individuals is sometimes critical.

The product which follows out of the implementation of this thesis, will be user-friendly

(no changing of batteries and no manual handling necessary), which gives it a huge

economic advantage. It can be reduced in size, so it does not bother the user and

makes the module easy to wear. You can’t expect people who require help, to change

batteries every now and then. This increases the autonomy and well-being of those

people. It also means that the created product would be more reliable than other exis-

ting products (no misplaced batteries or damage due to placing). The system sends

data signals on a regular basis so its status info is available (power level, out-of-range)

and if the signals are absent, an action can be triggered.

ENERGY HARVESTING 4

Chapter 3

Energy Harvesting

3.1 Solar

3.1.1 Introduction

The first source of energy which is examined is solar energy. Using photovoltaic cells

solar power is converted into electrical power. This method of energy scavenging is

already widely used and has proven itself to be viable for large and small scale purposes.

This makes solar energy a very promising source.

3.1.2 Solar module

Since the goal of this thesis is to make a compact transceiver, the solar panel must

be compact as well. Very small solar cells in an SOIC package are available, like the

CPC1824 [1]. Although these solar cells can be used to power a single IC, they don’t

produce enough power to drive a microcontroller and transceiver. About 100 µW is

what you can expect from the CPC1824 under optimum conditions. A better choice

is the SLMD121H10 IXOLAR SolarMD from the IXYS corporation [2]. This solar

module is considerably larger than the CPC1824 but, with a size of 42 by 35 mm, it is

still going to be smaller than the PCB. The solar module can be mounted on the back

to form a compact module.

A comparison between the two solar modules is shown in figure 3.1. The figure gives

the power density [mW/cm2] of the two solar modules as a function of load impedance

at an illuminance of 12 klux (measured in the shadow on a sunny day). It can be seen

that the SLMD121H10 delivers more power per unit of area.

3.1 Solar 5

Figure 3.1: Output power density of the CPC1824 and SLMD121H10 under different loadsat an illuminance of 12 klux

The power density curve of the larger solar module has its maximum power point at

a lower load impedance or equivalently, the larger solar module has a lower output

impedance. The reason becomes clear when you look at the larger solar module as

many smaller ones placed in parallel.

At optimal load and at an illuminance of 50 klux (direct sunlight, clear day) the

SLMD121H10 can provide 119 mW , the CPC1824 can only provide 1 mW. Because

of the inadequate output power of the CPC1824 the SLMD121H10 was chosen, but

there are a lot of suitable solar modules available. It is recommended however that the

chosen solar module can provide at least 20 mW under operation light conditions to

properly charge the batteries/supercapacitors that follow (cf. 4). The LTC3105 needs

a minimum input power to start operating. The minimum value of 20 mW (correspon-

ding with an illuminance of 10 klux) was experimentally determined.

The output impedance of a solar module becomes higher at decreasing light intensities

(cf. figure 3.3). At a certain light intensity the output impedance will become too high

for the following step-up converter to handle. If indoor solar energy harvesting (low

light intensity) is required, it is therefore better to use large solar modules with a low

3.1 Solar 6

open circuit voltage as they typically have a low output impedance. Here it is very

important that the open circuit voltage isn’t too low for the step-up converter that

follows (LTC3105).

Figure 3.2 shows the output power of the SLMD121H10 as a function of the load

resistance at different light intensities. It is clear that the maximum power point (the

output resistance) strongly depends on the light intensity. The output resistance of

the SLMD121H10 solar module is plotted in figure 3.3. To achieve maximum power

transfer at every light intensity, the regulator that follows should have a variable input

impedance. For further information on the SLMD121H10 the reader is referred to the

datasheet [2].

3.1.3 Regulator

Linear Technology has a wide variety of components specifically designed for energy

harvesting. This company is experienced in this sector. Their components have good

efficiencies and are used in a lot of different applications.

The LTC3105 [3] of Linear Technology is a step-up DC/DC converter specifically desi-

gned for solar energy harvesting from low voltage photovoltaic cells. Its input voltage

operating range is 225 mV to 5 V. An integrated maximum power point controller

allows for the user to set the operating point to achieve maximum power transfer from

the source. This is done by regulating the input voltage to a programmed value (via

a resistor on the MPPC pin). At a certain illuminance, the programmed voltage will

correspond with the solar module output voltage at the maximum power point. A

perfect match is only achieved at one illuminance level, but the voltage at maximum

power point does not vary much with light intensity. The voltage does strongly depend

on the used solar module so the LTC3105 can only be optimised for one solar module

at the time.

The schematic of the used solar harvesting circuit is shown in figure 3.4. A 3 F super-

capacitor load is placed at output of the LTC3105, this capacitor is used as the main

storage element (cf. section 4.3.2). The supercapacitor will be charged to a maximum

voltage of 3.7 V, this is set by the feedback resistors R1 and R2. Because the voltage

over the capacitor will vary, depending on the charge, another regulator is needed to

convert the voltage to a constant 3.3 V. Although this is not shown on figure 3.4, a

boost converter is intended. Since the intended boost converter (TPS61221) works

3.1 Solar 7

Figure 3.2: Output power of the SLMD121H10 as a function of load resistance at differentlight intensities

Figure 3.3: Output resistance of the SLMD121H10 solar module as a function of illuminance

3.1 Solar 8

with slightly higher input voltages than 3.3 V, a 3.7 V LTC3105 output voltage was

chosen to maximise the potential energy storage. The boost converter will be discussed

later in section 4.3.2.

A 300 kΩ resistor is placed at the MPPC pin of the LTC3105. This resistor sets the

minimum input voltage of the LTC3105 (the output voltage of the solar module) at

3 V. Since the efficiency of the LTC3105 depends on the input voltage (datasheet),

both the characteristics of the solar module and the characteristics of the LTC3105

should be taken into account when choosing this voltage. The 3V was chosen as an

optimum between the solar module efficiency and the LTC3105 efficiency. To take the

temperature dependency of the solar module into account, it is also possible to place

a thermally coupled diode on the MPPC pin. This was not done in this thesis.

Efficiency

The optimal input voltage of the LTC3105 depends on the output resistance of the

solar cell, which in turn depends on the light intensity and the temperature. The solar

cell characteristics aren’t the only parameters however, since the LTC3105 efficiency

isn’t the same for every input voltage.

Figure 3.5 shows the power efficiency of the LTC3105 under different output voltages.

This was measured at an illuminance of 65 klux (typical daylight illuminance, sunny

day). In this case Vin=2.6 V and Iin varied between 34.2 and 39.2 mA. This results in

an input power between 89 and 102 mW (cf. figure 3.6). The output was measured as

Figure 3.4: Schematic of the solar energy harvesting circuit

3.1 Solar 9

Figure 3.5: LTC3105 power efficiency under different output voltages at Vin=2.6 V (illumi-nance = 65 klux)

the 3 F supercapacitor load was charging. It can be seen that the efficiency strongly

depends on the output voltage. The reason is that the output current of the LTC3105

cannot exceed its input current. In the case of figure 3.5 the output current is 30 mA.

The LTC3105 already reaches this maximum current at a Vout of 1.2 V, therefore at

higher voltages the output power will increase linearly with Vout while the input power

varies only slightly between 89 and 102 mW (figure 3.6).

The power efficiency of the LTC3105 ranges from 21% at an output voltage of 0.7 V

to 89% at 3.4 V. However, the efficiency that matters is the total energy efficiency:

the ratio of stored energy on the supercapacitor to the total energy provided by the

solar module. Since the power efficiency is low at low supercapacitor voltages, the total

energy efficiency will depend on the voltage of the supercapacitor at the start of the

charge cycle (Vcap,start). When starting the charge cycle at a supercapacitor voltage of

0.7 V, the total energy efficiency over the whole capacitor charge cycle becomes 62.5%.

This charge efficiency becomes higher when charging the capacitor from a higher start

voltage (Vcap,start). This is shown in figure 3.7 where the total energy efficiency of a

charge cycle is plotted versus capacitor start voltage (Vcap,start).

3.1 Solar 10

Figure 3.6: Output and input power of the LTC3105 at an illuminance of 65 klux

Figure 3.7: Total energy efficiency of a charge cycle as a function of capacitor start voltage(Vcap,start)

3.1 Solar 11

To reach better efficiencies, the capacitor shouldn’t be discharged by the load too much

before the solar supply again becomes available. When starting the recharge cycle at

a Vcap,start voltage of 2.0 V for example, the energy efficiency becomes 79.9 %. This

results in an average power being stored on the supercapacitor of 77 mW. Figure 3.8

shows the average stored power as a function of Vcap,start at an illuminance of 65 klux.

A power of 77 mW is more than enough to power a pulsed sensor module.

Improvements

The output impedance of the solar module strongly depends on the light intensity (cf.

section 3.1.2). Therefore the operating point for maximum power transfer changes with

light intensity. In the current circuit this point is set by the resistor at the MPPC pin

of the LTC3105 and thus it cannot change with the light intensity. A possible impro-

vement might be to use a solar sensor to roughly measure the intensity. The maximum

power point might then be changed according to the light intensity. The CPC1824,

briefly mentioned in section 3.1.2, is suitable as a sensor. The solar cell is compact and

can easily be fitted onto the PCB. The sensor should be placed in parallel with a linea-

risation resistor and the output voltage should be measured with an ADC input of the

microcontroller (see section 5.3.3). The resistor value should be chosen as a trade-off

between resolution and linearity. A high resistor value increases the resolution since

the measured voltage (by the microcontroller) will cover a wider voltage range. The

resistance shouldn’t be to high or no distinction could be made between intensities at

higher intensity levels. A low resistor value increases the linearity because the number

of absorbed photons is proportional to the short circuit current.

3.1.4 Conclusion

Powers as high as 77 mW can be achieved in broad daylight (65 klux). This results in

about 7 mW/cm2. If this is compared with literature (e.g. [4] and [5]), this value is

comparable, but can still increase if the intensity is higher. This energy source will be

used to power the sensor module. The designed solar energy harvester is not designed

for indoor use. The light intensities are too low for our system to harvest any energy

from it. Other energy sources should be used.

3.2 Thermal 12

Figure 3.8: Average stored power as a function of the supercapacitor start voltage of thecharge cycle (Vcap,start) at an illuminance of 65 klux

3.2 Thermal

3.2.1 Introduction

Another energy form that can be harvested is thermal energy. Conversion from ther-

mal energy to electrical energy can happen via two distinct effects. In pyroelectricity a

voltage is generated by certain (pyroelectric) materials when heated or cooled. When

the material has reached a constant temperature, this voltage gradually disappears. In

thermoelectricity a voltage is generated by temperature differences. A thermoelectric

element creates a voltage difference when there is a temperature difference across the

element. This element also has the reverse operation. When a voltage is applied, both

sides of the element will be set at a different temperature. This can be used to cool or

heat objects.

The thermoelectric effect is best suited to harvest energy, especially for the purposes

of this thesis. One side of the element may experience heating, while the other side is

attached to a heatsink. The temperature difference will allow for thermal energy to be

harvested. One possible heat source is the human body, which is always available in

3.2 Thermal 13

personal applications. To generate power using the pyroelectric effect, a pyroelectric

element should constantly be heated and cooled. The number of applications which

use this effect are limited. Because of this, only the thermoelectric effect will be used.

The physics of the thermoelectric effect will not be discussed here but it is worth men-

tioning that the thermoelectric effect encompasses three different effects: the Seebeck

effect, the Peltier effect, and the Thomson effect [6]. The Seebeck effect is the conver-

sion of thermal energy to electrical energy. The peltier effect is effectively the opposite

i.e. the conversion of electrical energy to thermal energy. The Thomson effect handles

the heating or cooling of a wire with a temperature difference across its length and a

current flowing through it.

3.2.2 Peltier element

To generate power, using the thermoelectric effect, a peltier element will be used. As

mentioned before, one side of the element will experience heating while the other side

is attached to a heatsink so that this side remains at a more or less constant tem-

perature. The temperature difference will generate electrical energy (Seebeck). The

peltier element could of course also be used to generate a temperature difference when

a voltage is applied and thus using the Peltier effect rather than the Seebeck effect.

The used peltier element is the PolarTECTM PT4 from Laird Technologies [7]. This

is a compact (34 x 30 mm) thermoelectric module with maximum operating tempe-

rature of 80C, more than enough when using the human body as energy source. As

a first test, the power generated by the peltier element is measured (using different

resistive loads) as a function of the temperature difference. This results in figure 3.9.

It resembles an exponential function, but it will not continue increasing for higher

temperature differences due to temperature limitations of the peltier element. Remark

that these loads are not matched to the output impedance peltier element so they do

not provide the optimal power. In what follows, the optimal load will be determined.

Figure 3.10 shows the output power as a function of load impedance for the PolarTECTM

PT4 at a temperature difference of 20C. From the figure it follows that the internal

resistance of the peltier element is roughly 8 Ω and that, at a temperature difference of

20C, the maximum output power is 3.6 mW. Because of the low output impedance,

the generated voltage is low (173 mV at maximum power point but generally much

lower). A converter is needed to boost the voltage to 3.3 V, this converter has to be

3.2 Thermal 14

Figure 3.9: Output power of the PolarTECTM PT4 as a function of the temperature diffe-rence at different loads (1.2 kΩ and 12 kΩ)

Figure 3.10: The Output power as a function of load impedance for the PolarTECTM PT4at a temperature difference of 20C

3.2 Thermal 15

able to operate at extremely low input voltages.

3.2.3 Regulator

A chip from Linear Technology (cf. section 3.1.3) will also be used for the thermal

energy harvesting chip.

The LTC3109 [8] is a DC/DC converter designed for very low input voltages. The

input voltage operating range goes from 30 mV to 500 mV. The LTC3109 has an auto-

polarity function which allows for energy harvesting regardless of polarity of the input

voltage. To this end, the converter uses two external step-up transformers.

The LTC3109 uses internal switches to form an oscillator and the AC signal produ-

ced is then boosted and rectified internally. The minimum input operating voltage is

determined by the turn ratio of the external step-up transformers. A ratio of 1:100 is

recommended to reach voltages as low as 30 mV. If a higher minimum input operating

voltage is allowed, a lower transformer ratio is a better choice since the efficiency is

higher at these lower transformer ratios.

If the polarity in the intended application doesn’t change, de LTC3108 [9] could be

used. This regulator is almost exactly the same but has no autopolarity function and

the efficiency is slightly higher. Both the LTC3109 and LTC3108 have been specifically

designed for thermoelectric energy harvesting.

Schematic

The used schematic for the LTC3109 thermoelectric energy harvesting circuit is shown

in figure 3.11. The output voltage (Vout) is 3.3 V but this can be changed to 2.35 V,

4.1 V or 5 V. When the output has reached regulation at 3.3 V, the incoming energy

is stored on the 0.33 F supercapacitor at the Vstore pin. The supercapacitor is char-

ged to 5 V. When the input source becomes unavailable, the energy stored on the

supercapacitor will be used as backup for the output but only when the voltage on the

supercapacitor is larger than the programmed output voltage (in this case 3.3 V). This

is done by an internal buck converter.

A trade-off has to be made between a larger and a smaller supercapacitor value. A

larger capacitor is able to store more energy, but is completely useless when the su-

3.2 Thermal 16

percapacitor voltage never becomes larger than 3.3 V. A smaller capacitor is charged

more quickly to a voltage > 3.3 V and can thus be used more quickly as backup but

the capacitor can store less energy. In the schematic a value of 0.33 F was chosen but

the ideal value will depend on how much thermal energy is expected and on how long

the thermal source will become unavailable.

The LTC3109 has a few other pins which aren’t used in this design. The Vout2 pin is

exactly the same as Vout, only the output can be disabled by the Vout2 EN pin. This

can be used to shut down sensors who don’t have a sleep mode. The PGOOD pin is

an indicator for when the output has reached its programmed voltage. Lastly, a 2.2

V (internal) low dropout regulator (LDO) is available, but it is not used since none of

our intended components will work at a 2.2 V supply voltage.

Efficiency

Figure 3.12 shows the LTC3109 power efficiency under different input voltages at a

load of 12kΩ. The load was chosen so that at no tested input voltage enough power

would be transferred to the load for the output voltage to reach 3.3V. Since the load

doesn’t reach 3.3 V (in the test case Vout,max was 2.4 V) no energy will be transferred

to the store pin and all the available energy will go to the load. This allows for an

easier efficiency measurement.

As said before, the used peltier element has a very low output impedance (8 Ω) and to

measure the input power, the input current has to be measured. The internal resistance

of the available current meters was too high and resulted in incorrect measurements.

Instead the voltage across a 220 mΩ series resistance was measured.

As seen in de efficiency plot, the maximum efficiency is 28.3 % at an input voltage of

80 mV. This is low in comparison with regular DC/DC converters but regular DC/DC

converters don’t have a minimum input operating voltage as low as 30 mV. Higher

efficiencies are useless if the used converter cannot work at a certain input voltage. A

choice had to be made between higher efficiencies and lower minimum input operating

voltage. The voltages produced by the peltier element when using a human body as a

heat source are not spectacular (< 100 mV).

The power gathered from the peltier element when using body heat (the plates of the

3.2 Thermal 17

peltier element had a temperature difference of 10C) is 480 µW . With the used peltier

element measuring 3 by 3.4 cm, this results in a power density of 47 µW/cm2.

3.2.4 Conclusion

The results which were obtained in this section 3.2.3, can be compared with literature

([4] and [5]). The gathered power depends on the actual temperature difference but

with a power density of 47 µW/cm2 the results are in the same order of magnitude

as found in literature. For an outdoor environment (sunny day) the obtained power is

lower than when using solar energy. However for an indoor environment more energy

can be gathered from thermal energy than from solar energy. Thermal energy will be

used in our design to power the sensor module.

Figure 3.11: Schematic of the thermal energy harvesting circuit

3.3 Vibration 18

Figure 3.12: LTC3109 power efficiency under different input voltages at a load of 12kΩ

3.3 Vibration

3.3.1 Introduction

Vibration is the third source for EH, which is examined. In the application, the sensor

is used as a personal transceiver, which means that it is carried along with a person.

Movement vibrations are thus frequently present. This makes vibration an interesting

source to investigate.

To harvest this energy, a piezoelectric element is used. Such an element generates an

AC voltage with a frequency equal to the vibration frequency. This voltage has to be

rectified and converted to a usable DC voltage source.

3.3.2 Piezoelectric element

The piezo element V22BL from Volture [10] is especially made for energy harvesting

(figure 3.13, dimensions: 6.35 cm × 0.61 cm). It can produce high voltages when

triggered. This will make the rectifying step much more efficient. The voltage drop

over the diodes will be relatively small compared to the input voltages. The output

impedance of the element is very high, which makes the output current very small.

3.3 Vibration 19

There are two piezoelectric wafers on the V22BL. They can be connected in series or

in parallel to respectively double the voltage or the current. Another option, which is

determined according to the application is tuning the Volture piezo energy harvester.

This can be done by adding a tip mass on the edge of the cantilever. This will change

the resonance frequency to a frequency which is common in the application if the mass

is designed and chosen correctly. How to choose the mass can be determined on page

13 of the datasheet [10].

3.3.3 Full wave rectifier and buck converter

The LTC3588-1 of Linear Technologies [11] will be used. This chip will convert the AC

voltage of the V22BL to a usable DC voltage. It is specially made for converting the

energy of piezoelectric elements with a high output impedance. It consist of both the

full wave rectifier and a buck converter.

Internally the chip first rectifies the output of the piezoelectric element using a standard

4 diode full wave rectifier. The DC power generated is stored on a large external

capacitor, which can reach 20 V (limited by an internal zener diode). At the start, the

voltage on the storage capacitor will rise until it reaches 5 V. The output voltage stays

zero. At 5 V, a part of the charge on the storage capacitor (Vin) will be transferred to

Vout. Vin has dropped and will recharge to 5 V using the energy from the piezoelectric

element. At 5 V, the same charge transfer will happen. This cycle will repeat itself

until the preset value of Vout is reached. Then the storage capacitor will keep charging

until 20 V is reached. The created circuit in shown in figure 3.14. The output voltage

is set to 3.3 V (D1 = Vin2 and D2 = 0, with Vin2 an internal low voltage rail at 6 V).

Some rudimentary measurements were done. The functionality was tested to see if

the voltage at Vin dropped when it exceeded 5 V. Some charge is transferred to the

output capacitance. The efficiency of the first step (drop of Vin) can be calculated by

examining the voltage drop at the storage capacitor and the voltage step at the output.

efficiency =

CoutV 2out,1

2− CoutV 2

out,2

2CinV 2

in,2

2− CinV 2

in,1

2

= 68%. (3.1)

The efficiency of the rectifier is not measured. Some test are performed using the series

configuration of the piezoelectric elements and some with the parallel configuration.

The main difference is that when the current is doubled (parallel configuration) the

output voltage reached regulation more quickly.

An important remark is that the capacitor at the input discharges quite quickly, e.g.

3.3 Vibration 20

Figure 3.13: Piezoelectric element: V22BL

Figure 3.14: Schematic of the piezoelectric energy harvesting circuit

3.4 RF 21

2.4 µW at 3.5 V.

3.3.4 Conclusion

A better insight is achieved in vibration energy, but it became clear that the power

harvested would be much smaller in comparison to the first two sources of energy

harvesting. In literature [4], this anticipation was confirmed with power densities of 4

µW/cm2. Therefore, vibrational energy will not be used to power our transceiver.

3.4 RF

3.4.1 Introduction

More and more communication is performed wirelessly. This means that more and

more ambient RF energy is available due to billions of RF transmitters around the

world. These transmitters are e.g. personal (private) devices, mobile base stations and

broadcast stations. The amount of wireless data transfer and thus energy will only in-

crease in the next years. This makes energy scavenging using RF energy an interesting

option to investigate.

The first component to intercept RF energy is an antenna. The AC voltages induced

onto the antenna have to be converted to a useful DC voltage. For this, a rectifier

is needed. A general name for the combination of these two components is rectenna.

Optionally, some pre- and post-rectifying filters can be added. Like in the previous

sections (3.1, 3.2 and 3.3) the obtained voltages have to be boosted using a step-up

converter. The output from the converter can be used to charge a supercapacitor (cf.

chapter 4).

3.4.2 Possibilities

There are a lot of possibilities on how the energy can be harvested determined by the

frequency band in which one can harvest easily and with the highest energy levels, and

the choice of antenna shape and material. These choices are discussed below.

Frequency

The RF-band (3 kHz - 300 GHz) consists of many frequencies that can be exploited

to extract energy. At high frequencies (2.4 GHz and 5 GHz) Wifi is promising. It is

3.4 RF 22

widely spread and commonly used but a wireless router does not emit a lot of power

(maximum 20 dBm). Because of this, using Wifi is viable for domestic use, close to

the transmitter. For long range operation a high power transmit station is needed like

a mobile/TV/Radio base station. Above 1 GHz these are not commonly available.

The spectrum can be visualised by a spectrum analyser and an adjustable monopole

antenna SRH789 [13] from the lab. This antenna can be altered to resonate at frequen-

cies between 95 MHz and 1100 MHz. Not every frequency has the same gain when

the antenna is set to the correct length, but a general overview of the interesting and

powerful frequencies can be deduced. The gain of the monopole fluctuates between

2.15 dBi and 3.2 dBi when an optimal length is used. For the lower frequencies (95

MHz to 300 MHz), the antenna acts as a quarter wavelength monopole, while at higher

frequencies (300 MHz - 1100 MHz) the antenna has a length of 58

of the wavelength

like indicated on the antenna.

The local spectrum [10 MHz, 990 MHz] is given in figure 3.15 (resolution bandwidth

(BW) is 3 MHz, averaged over 100 samples). The power values indicated on this graph

are not the actual optimal powers. The antenna is set to a length of 44 cm which

means that it is only tuned for 2 frequencies but not for the entire spectrum. When

measuring a smaller frequency band (see later) the antenna can be set to a more ap-

propriate length. The origin of every large peak is indicated on the spectrum. Note

that the power on the graphs is the power integrated over 1 resolution BW.

Now every large peak will be discussed separately. The channel power from each of

these channels is measured using the MXA signal analyser N9020A from the lab. The

data is averaged over 100 samples and now a resolution BW of 100 kHz is chosen.

The indicated powers are now the result of an integration over 100 KHz (the BW).

The values of the powers below are highly dependent on the exact location and time

at which the measurements are done. The mutual relationship between the channel

powers is less variable so a rough estimate of the most promising channels can be made.

If integrated over the entire peak, the channel power is obtained.

• FM-Radio: On figure 3.16 the commonly used FM-Radio channel is displayed.

It ranges from 87.5 MHz to 108 MHz. The wavelength is about 3 meter and its

channel power is -33.5 dBm.

• T-DAB: The second major channel is a Terrestrial-Digital Audio Broadcast (T-

DAB) channel (figure 3.17) which ranges from 223 MHz to 224.8 MHz. The

3.4 RF 23

Figure 3.15: Spectrum below 1 GHz, Ghent

power in this channel is about -39.5 dBm and has a wavelength of about 1.33 m.

• DVB-T: In the next two figures (3.18 and 3.19) the effect of Terrestrial Digital Vi-

deo Broadcasting is shown in different frequency bands. The channel powers are

respectively -42.5 dBm and -39.5 dBm (sum of three channels) with wavelengths

of 0.62 m and 0.46 m (average of three channels).

• GSM: The last band which can be used to harvest energy is the well known 900

MHz GSM-band. The most activity was seen in the band between 900 MHz and

960 MHz. The peaks vary a lot and the channel power is fluctuating between -42

dBm and -36 dBm.

Antenna choice

There is a wide variety of antennas available. The most important criteria to choose

an antenna for energy harvesting are summarised below. These criteria are specifically

for the application of this thesis.

• Radiation pattern: If the radiation which has to be intercepted, does not come

from a certain direction (or the antenna does not have a fixed orientation) the

antenna should be rather omnidirectional to harvest enough energy in every direc-

tion. If a certain direction contains more radiation the radiation pattern should

3.4 RF 24

Figure 3.16: Spectrum FM-Radio, Span35 MHz

Figure 3.17: Spectrum T-DAB, Span 35MHz

Figure 3.18: Spectrum DVB-T at 482MHz, Span 35 MHz

Figure 3.19: Spectrum DVB-T at 650MHz, Span 70 MHz

have maximum gain towards that angle, but this is not the case for a portable an-

tenna with no fixed orientation. The omnidirectional gain of the antenna should

be as high as possible.

• Impedance bandwidth: A higher bandwidth will allow more frequencies to contri-

bute to the harvested energy.

• Integration: It should not pose a problem to integrate the antenna into our sensor

module without occupying too much space.

The kind of antenna implemented on a PCB can easily be integrated (in the application

2). More specific, a microstrip patch antenna. At the front side of the patch antenna

the gain is much higher than at the back side. This means that the patch antenna

should be placed with its front side to wherever the radiation is coming from. This is

not always possible due to the portability of the antenna. Important advantages are,

that the antenna has a low profile, a small cost and it is easy to fabricate. The total

size of the system has to be kept small. If the antenna would be integrated into the

3.4 RF 25

Figure 3.20: Spectrum GSM, Span 70 MHz

system, it would be situated on the bottom plane of a PCB. The circuit that converts

the harvested energy into a usable DC voltage can also be placed on the same PCB

but should have a small impact on the antenna performance. An interruption of the

ground plane should be avoided.

Matching and dimensions

When energy is harvested using an antenna, this antenna has to be matched to the

follow-up system. There are two possibilities of impedance matching [14]:

• Lumped element: Using resistors, capacitances and inductors, the antenna can be

matched to the follow-up system. This way optimal power transfer is guaranteed

for a certain frequency or frequency band.

• Distributed: It is also possible by using structural modifications of the antenna

layout, stubs, transmission lines, dielectric loading etc. to match the circuit.

A mix of both techniques can be used. One will start with an antenna which is matched

for certain frequencies and resonates. This way the input impedance is purely resistive

and all the available power is absorbed by the antenna. If needed, extra matching (if

not perfectly matched) can be added using lumped components and transmission lines

(stubs).

To dimension a simple rectangular patch antenna which resonates at a certain frequency

f, the following formula (3.2) can be used. It gives an approximation of the length of

the patch [15].

3.4 RF 26

L =λ

2√εr

(3.2)

In this formula, λ is the free space wavelength and εr is the relative permittivity of

the substrate. This is a rude approximation, but can be used as a guideline. This

formula assumes that all the fields are contained within the substrate. This is not

the case because of fringe fields (which cause radiation) partially situated in the air.

This causes the effective relative permittivity εr,eff to be somewhere in between 1 and

εr. Therefore the length will have to be larger than predicted by formula (3.2). The

resonance frequency can be determined by simulations or experimental formulas like

e.g. formula (3.3) from literature [16].

εr,eff ≈εr + 1

2+εr − 1

2

[1 +

10h

W

]− 12

(3.3)

L =λ

2√εr,eff

(3.4)

In this formula, h is the height and W is the width of the patch antenna.

If a low frequency band (with a large wavelength) has a lot of channel power, a patch

with large dimensions is needed to capture this energy efficiently. This does not fit

with the purpose of the thesis. A compact antenna is needed. It is possible to keep

the dimensions small by increasing the relative permittivity of the dielectric. The most

frequently used dielectric for PCBs is FR4 which has a relative permittivity of 4.25. It

is also possible to use other PCB materials to fabricate the patch e.g. RO3010 from Ro-

gers Corporation [12]. RO3010 is a high frequency PCB material which has a εr of 11.2.

E.g. if a patch antenna has to be made for the T-DAB (224 MHz) band, the dimensions

should be at least in the order of 32 cm when using FR4 as dielectric. This is far to

large for a compact design. If RO3010 is used, this becomes 20 cm, which is still large

but this can be incorporated more easily into e.g. clothing, package.

The amount of power transmitted (received) does also strongly depend on the thickness

of the substrate. Transmitted (or transceived) waves all origin from the fringe fields at

the side of the patch antenna (cf. figure 3.21). Therefore a thicker substrate is able to

emit more fringe fields than a thin substrate due to the larger side surface.

3.4 RF 27

Fringe fields

Substrate

Patch

GND

Substrate

Figure 3.21: Patch antenna, fringe fields

3.4.3 Simulations and measurements

Using ADS (Advanced Design System [17]), the RO3010 substrate is simulated for

some simple patch antennas. The first patch has a substrate thickness of 1.28 mm, the

second one 0.64 mm. They are dimensioned using the simple formula (3.2) at a center

frequency of 940 MHz. This frequency lies in the middle of the GSM band (cf. figure

3.20). A 1 GHz FR4 patch antenna was already available in the lab. It has a thick-

ness of approximately 0.83 mm. Figure 3.22 assures that (for FR4) the simulation and

measurements coincide fairly well. ADS simulates the ground and parameters correctly.

On figure 3.23, the S-parameters of the simulated RO3010 antennas are displayed. At

resonance, the thin substrate reflects less power, due to a better matching (same width

of the line feed, while substrate thickness is different). The resonance frequency also

slightly changes due to the difference in thickness of the substrate. In figure 3.26, the

actual measurements on the fabricated antenna are shown. The same conclusion can

be made about the reflection and resonance frequency.

Figure 3.22: FR4 patch: S(6,6) simulation, S(5,5) measurement

3.4 RF 28

In the next two figures (figure 3.24 and 3.25), the difference between the simulations

and the measured values are plotted. Note that the measurements have a higher re-

sonance frequency and a different shape. The reason that the curves do not coincide

is a bad setting of the parameters of the substrate (RO3010). Using formula 3.4, the

effective permittivity can be determined (L = 50 mm). Using this εr,eff and formula

3.3, the relative permittivity becomes 10.08 and 10.35 for respectively a thickness h of

0.64 mm and 1.28 mm. If those values for εr are inserted into ADS, they give rise to

a resonance frequency which is too high.

When tweaking the relative permittivity in ADS to make the resonance frequency from

the simulation and measurement coincide, a value of 10.75 is obtained for both thick-

nesses. This characterisation is summarised in table 3.1.

The matching and dimensioning of the antennas on the SO3010 is not very good be-

cause the feed lines are to wide (2 mm), the width and injection point are poorly

dimensioned. It would be better to use a line of 0.51 mm for the thin substrate and

1.05 mm for the thick substrate. This was calculated using linecalc, a tool in ADS. To

choose the other dimensions, some literature was used ([18] and [16]). Simulating this

patch, a general idea can be formed of what can be harvested using SO3010. Note that

the tweaked parameters for SO3010 are used. The results are shown in figure 3.27.

Remark that optimisation leads to very high Q-factors that rapidly change with small

modifications of the topology. So the value at resonance is not very representative for

the real situation.

Figure 3.23: Simulation RO3010 patches: S(2,2) 1.28 mm, S(4,4) 0.64

3.4 RF 29

Figure 3.24: SO3010 patch (1.28 mm): S(4,4) simulation, S(3,3) measurement

Figure 3.25: SO3010 patch (0.64 mm): S(2,2) simulation, S(1,1) measurement

Figure 3.26: Measurement SO3010 patches: S(3,3) 1.28 mm, S(1,1) 0.64 mm

3.4 RF 30

Table 3.1: Characterisation SO3010 substrate permittivity

Parameter Datasheet Experimental formula ADS Tweaking

εr 11.2 10.08-10.35 10.75

Figure 3.27: Simulation optimised SO3010 patches: S(1,1) 1.28 mm, S(2,2) 0.64 mm

The main problem when taking a material with a larger permittivity is the decreased

energy transfer from air to the substrate. Using formula (3.5) from the course Applied

Electromagnetism [19] the amplitude transmission coefficient can be calculated.

Tn =2

1 + Z0

Zs

(3.5)

With Z0 and Zs respectively equal to the characteristic impedance of the air and the

substrate. To get the characteristic impedances of the substrate, formula (3.6) is used.

Zs =

õ

ε

1√1 + j · tan(δ)

(3.6)

Note that the last factor is removed when losses can be neglected. This happens

when the loss tangent (tan(δ)) is much smaller than 1. This is the case for both FR4

(tan(δ) ≈ 0.018) and RO3010 (tan(δ) = 0.0022 − 0.0011). The permeability of both

the materials is approximately equal to the permeability of the air. This results in:

Z0 ≈√µ0

ε0(3.7)

ZFR4 ≈√

µ0

4.25 · ε0(3.8)

3.4 RF 31

ZRO3010 ≈√

µ0

11.2 · ε0(3.9)

Using (3.7),(3.8) and (3.9) it is possible to calculate the amplitude transmission coef-

ficients:

TFR4 =2

1 +√

4.25= 0.65 (3.10)

TRO3010 =2

1 +√

11.2= 0.46 (3.11)

This means that, using a larger dielectric constant, less power will enter the substrate

(so less power harvested). Using formula (3.12), twice as much power is available using

FR4. Note that this comparison is for patches which have the same surface and incident

power. The frequency of the incident signal has to be tuned for both the patches.

Power Ratio =0.652

0.462= 2.00 (3.12)

3.4.4 Conclusion

In the previous sections, a general idea and insights about RF energy harvesting were

created. It is definitely possible to use RF energy harvesting to power some sort of

device (like many examples on the internet proof). On the other hand, there are energy

sources with a lot more power. The average power harvested using ambient RF waves

is in the order of magnitude of 0.1-1 µW/cm2 ([4] and [5]). This is much less, than

what can be (and is) achieved using solar power and heat (one hundred to one hundred

thousand times more power). In this thesis, no module (rectifier and step-up converter)

will be designed that harvests energy out of RF power.

There are exceptions and conditions where RF energy harvesting is preferable/viable:

• Freedom of space: If more space is available (than in the suggested application)

more power can be received. More space opens up more possibilities concerning

different kinds of antennas e.g. loop antennas. These loop antennas can be made

resonant by placing a capacitor at the base.

• Freedom of thickness: A thicker substrate would also improve the amount of

power that is transferred to the harvesting circuit. Very thick substrates are not

commonly available.

3.4 RF 32

• Low frequencies: If a patch antenna is used, frequencies below 700 MHz are not

suitable to harvest due to size limitations. Enlarging the εr is not an optimal

solution if the power at low frequencies is comparable to the power at high fre-

quencies. If the low frequency power is a factor two higher (cf. (3.12)), using

RO3010 can become a viable solution. If a compact design is required, different

kinds of antennas can also be used to capture low frequencies.

• High power source: If a high power external source of RF energy is present (can

be self-generated) the energy that can be harvested is more significant compared

to the other sources.

• Low transmit frequency: If the transceiver (which is powered by the energy har-

vesting) sends very rarely, maybe enough energy can be collected in the interval

between two transmits to withstand the sudden burst of needed energy.

STORAGE 33

Chapter 4

Storage

4.1 Introduction

In the previous chapter energy harvesting was discussed. Since it is possible and likely

that more energy will be harvested than can be consumed at a given time, storage

elements have to be added as not to waste energy. Moreover in applications where the

energy source is not available all the time (e.g. solar applications), enough energy has

to be stored so that normal operation isn’t disrupted when the energy source becomes

unavailable.

Possible storage methods are conventional batteries (coin cell, NiMH, Li-ion, ...), thin-

film batteries or supercapacitors. The advantages and disadvantages of these different

types of storage elements will be discussed in the next section. After that the imple-

mentation will be discussed.

4.2 Storage methods

In this section an overview of the different available storage devices is given. The

information was gathered from various sources ([20], [21], [22], [23], [24], [25], [26]).

4.2.1 Conventional rechargeable batteries

The most obvious energy storage devices are the conventional rechargeable (electroche-

mical) batteries such as NiCd, NiMH, and Li-ion batteries. Each battery type has its

own advantages and disadvantages and a comparison between them will not be given

here. The most important advantages and disadvantages of electrochemical batteries in

4.2 Storage methods 34

general (in comparison with supercapacitors and thin-film batteries) are given below.

Advantages

• Flat voltage profile. Conventional batteries have an output voltage that more

or less remains constant for the most part of their charge cycle. The battery

voltage only starts to drop dramatically when up to 90% of the energy has been

discharged. This is a clear advantage with respect to supercapacitors.

• High energy density. Electrochemical batteries can have more than 10 times the

energy density of supercapacitors. Only fuel cells have higher energy densities.

Disadvantages

• Low number of recharge cycles. Rechargeable electrochemical batteries are infe-

rior to non-rechargeable batteries (lower energy densities, higher self-discharge,

...). Their electrical characteristics only get worse after each recharge cycle. The

most important changes are a decrease in energy capacity and a decrease in out-

put voltage. Rechargeable coin cells can only be recharged several 100 times

before their characteristics become inadequate.

• Self-discharge. Rechargeable electrochemical batteries have a higher self-discharge

than non-rechargeable batteries, the battery may be discharged in a week. This is

better than supercapacitors but much worse than solid-state thin-film batteries.

• Low power density. Electrochemical batteries have a much lower power density

than supercapacitors. Because of this, they are not suited for pulse current

applications.

• Slow charging. The charging of an electrochemical battery can take up to several

hours. Supercapacitors can be charged in several minutes.

• Affected by complete discharges. Electrochemical batteries deteriorate quickly

when they are discharged completely. It is very important that the battery

voltage doesn’t drop below a certain threshold.

• High environmental impact. Electrochemical batteries and especially rechar-

geable electrochemical batteries contain many toxic chemicals. When the batte-

ries are thrown away or begin to leak they can contaminate surface or ground-

water supplies. Lithium is used as a psychiatric drug and has an effect on the

human brain. Care must be taken when disposing of used batteries.

4.2 Storage methods 35

• Flammable. Many of the battery chemicals, especially lithium, are flammable.

When the battery combusts, the damage may not be restricted to the battery

itself. The electrical circuit may be destroyed and poisonous fumes may be re-

leased.

• Large physical size. In comparison with solid-state thin-film batteries, electro-

chemical batteries are bulky and heavy.

4.2.2 Supercapacitors

Supercapacitors, ultracapacitors or electric double-layer capacitors (EDLC) are capa-

citors with an extremely large capacitance value for their size and weight. They can

have an energy density 10 to 100 times higher than conventional elektrolytic capacitors.

In general, they have a bad frequency response but an excellent pulse response [22].

Because of this they are suited to store energy and deliver current pulses to a load.

The construction or the internal composition of supercapacitors will not be discussed

here but the main advantages and disadvantages are summarised below.

Advantages

• High output power. Supercapacitors have a power density that can be 10 to 100

times higher as conventional batteries. Because of this, they are suited for pulse

current applications where high and short current pulses are needed.

• Fast charge. While conventional batteries can take up to several hours to re-

charge, supercapacitors can be charged in several minutes or lower. This is

because they don’t rely on electrochemical energy storage but physical charge

storage. Supercapacitors also don’t need to be charged with a constant current

and they can handle high current peaks.

• Low impedance. Supercapacitors have a low output/input impedance which al-

lows them to handle current peaks. They can also be placed in parallel with a

battery to improve load handling. High current peaks will be provided by the su-

percapacitor while the battery provides more storage. This can increase battery

life significantly.

• Long life. Supercapacitors last much longer than conventional batteries. While

batteries deteriorate when constantly being charged and discharged, supercapaci-

tor life is mostly just a function of time at a certain temperature and voltage [22].

There is little degradation over hundreds of thousands of charge cycles (much hi-

gher than rechargeable batteries). The service life of supercapacitors can reach

4.2 Storage methods 36

10 to 15 years which means that they often will outlast the rest of the circuit.

This can significantly reduce maintenance costs.

• Not affected by deep discharges. Electrochemical batteries deteriorate quickly

when they are discharged completely, this is not the case with supercapacitors.

• Improved safety. Supercapacitors are not flammable and won’t explode.

• Low toxicity. Supercapacitors do not release any hazardous substances that can

damage the environment, unlike many batteries.

• Increase in capacitance when the temperature decreases below the rating tempe-

rature. This is not the case for electrochemical batteries.

Disadvantages

• Low energy density. Supercapacitors have energy densities that are approxima-

tely 10 times lower than conventional batteries.

• High self discharge. The self-discharge rate is higher than that of an electroche-

mical battery.

• Rapid voltage drop. As in any capacitor the voltage drops linearly with the total

charge on the capacitor. A regulator will be needed to supply a constant voltage.

This will introduce extra losses.

• More expensive. When comparing supercapacitors with conventional batteries

of the same capacity, supercapacitors are more expensive. ”Since the electro-

static device life of the supercapacitor is much longer than the service life of

electrochemical batteries, they often have a lower long term total cost.” [27]

• Initial capacitance fade. The capacitance value can quickly drop in the beginning

of its service life. The capacitance value continus to drop (more slowly) during

the rest of the service life. The internal resistance also slowly increases during

the service life. These ageing effects have to be taken into account when selecting

the supercapacitor.

• Low maximum voltage. Supercapacitors cannot handle high voltages. A series

connection of more than 1 supercapacitor is needed to obtain higher voltages.

Voltage balancing circuits are recommended when 2 capacitors are placed in series

and required if more than three capacitors are connected in series. This is because

no 2 components are the same and 2 supercapacitors can have a different leakage

4.2 Storage methods 37

current (at the same voltage). Since in a series connection the leakage current of

1 supercapacitor must flow in the other, the voltage over the capacitors will not

be equally divided (one capacitor will have a higher voltage than the other so

that their leakage currents are the same). So without proper balancing circuits

it may still be possible that the maximum ratings are exceeded.

• Integration issues. The supercapacitors are derated in capacity when soldered at

high temperatures. Reflow soldering is not recommended.

4.2.3 Thin-film batteries

Unlike conventional batteries, thin film batteries (also called laminar batteries) are

composed of solid-state thin films. They contain lithium but in a much smaller dose

than conventional batteries. They can have a thickness of 50 µm and in many cases they

can be made flexible. Thin-film batteries are available from companies as Excellatron,

Infinite Power Solutions, Cymbet or Solicore. They can be made on silicon wafers using

semiconductor processes. The main advantages and disadvantages are summarised

below.

Advantages

• Flat voltage profile. As with conventional batteries, the output voltage of thin-

film batteries more or less remains constant during discharge. The voltage only

starts to drop rapidly when 80% of the energy has already been discharged.

• Very low self-discharge. Of all the storage devices discussed here, thin-film bat-

teries have by far the lowest self-discharge. There is an energy loss of about 1 or

2 % per year as opposed to 10 to 20% per day for supercapacitors [26].

• Minimal device ageing. Conventional rechargeable batteries (especially coin-cells)

are notorious for quickly losing their capacity when recharging too many times.

Thin-film batteries like the cymbet CBC050 still retain 80% of the specified

capacity after 5000 recharge cycles.

• Fast charging. Thin-film batteries can charge faster than conventional batte-

ries, the cymbet CBC050 charges to 80% of total charge in under 20 minutes.

Conventional batteries can take up to several hours.

• Small footprint. Thin-film batteries have a more convenient size than electro-

chemical batteries or supercapacitors. They can be packaged in a conventional

ic-package. Another possibility is co-packaging. The thin-film battery may be

4.2 Storage methods 38

packaged together with other circuits such as real-time clock devices. This result

in lower losses and an even smaller physical size.

• Flexible. Some thin-film batteries can be made very flexible, they can be bend

and twisted without damage.

• Easily integrated. Because of their small size and conventional packaging, they

can be easily integrated. Reflow soldering is allowed.

• Environment friendly. Thin-film batteries have a very small chance of combustion

and they consists of less lithium, improving environmental credentials and health

benefits (because of the effect lithium has on the brain). All of this results in a

reduction in end of life cost [20].

• Damage tolerant. Thin-film batteries are solid-state devices. They can handle

more physical abuse than conventional batteries.

Disadvantages

• Limited charge current. Just as with conventional batteries, the charge current

is limited. Thin-film batteries can be charged faster than conventional batteries

but not nearly as fast as supercapacitors, which are not current limited.

• Limited recharge cycles. Although thin-film batteries can be recharged much

more than conventional batteries, they still have a disadvantage over supercapa-

citors in this regard.

• Limited capacity. Although thin-film batteries have higher energy densities than

conventional batteries, there are no high-capacity thin-film batteries available.

Several thin-film batteries can be combined and they will have a lower weight,

but this quickly becomes impractical and very expensive.

4.2.4 Choice of the storage component

A comparison between supercapacitors and conventional batteries with regard to their

energy and power densities can easily be seen in a Ragone chart (Figure 4.1 [28]), this

chart plots energy density as a function of power density. It can be seen that conven-

tional electrochemical batteries are better for large energy storage. Thin-film batteries

aren’t shown in the chart but they can have an even higher energy density while still

having a decent power density. For the time being however, thin-film batteries aren’t

4.3 Implementation 39

Figure 4.1: Ragone chart [28]

produced with large storage capacity. Combining multiple thin-film devices to obtain

a higher energy capacity can quickly become very expensive.

Supercapacitors have a lower energy density (10 times lower) than conventional batte-

ries but they can have a power density 10 to 100 times higher, making them ideal to

provide high current pulses to a load.

As seen in the chart, fuel cells have a higher energy density than conventional batteries.

Because of their low ignition point they were never considered as a suitable storage

device.

Even though conventional batteries have a higher energy density than supercapacitors,

because of their short life span and higher environmental impact the choice was made

to use only supercapacitors and thin-film batteries as storage elements.

4.3 Implementation

4.3.1 Thin-film battery

Thin-film batteries are available from companies like Excellatron, Infinite Power Solu-

tions, Cymbet or Solicore. The choice was made for the EnerChips of Cymbet because

of their availability and conventional packaging.

4.3 Implementation 40

Figure 4.2: Typical EnerChip discharge characteristic as copied from datasheet CBC3150[29]

Cymbet Enerchip

The Cymbet corporation produces thin-film batteries [30] and these batteries come in

two forms:

The EnerChip CBC050 [31] is a solid state rechargeable battery with a capacity of

50 µAh (at 3.8 V). As with all conventional batteries the capacity depends on the

discharge rate as depicted in figure 4.2. The rated capacity of 50 µAh is at a discharge

rate of 100 µA. As with conventional batteries the capacity of the battery depends on

the discharge current. At a higher discharge rate the capacity will be lower and at a

lower discharge rate the capacity will be higher than the nominal capacity.

The EnerChip CC CBC3150 [29] consists of the same battery but also includes a circuit

for controlled charging of the battery and has a separate output which can be adjusted

and thus differ from the battery voltage. The chip has a built-in energy storage pro-

tection, making sure that the internal battery isn’t overcharged. The internal charge

pump charges the battery from input voltages ranging from 2.5 to 5.5V without the

need for an external control circuit.

The EnerChip CC CBC3150 also acts as uninterruptible power source. When the input

source is available (and between 2.5 and 5.5V), the input is connected to the output of

the device. However, when the source becomes unavailable the output voltage is set at

the user-defined voltage (between 2.5 and 5.5V) with the energy being provided by the

4.3 Implementation 41

battery. The battery voltage is relatively constant and varies between 4.1V and 3.8V

for the most part of its charge cycle. As with most solid state batteries the device has a

low self-discharge and the battery almost doesn’t deteriorate, this allows for thousands

of recharge cycles. After 5000 recharge cycles (at 10% depth of charge) the EnerChip

retains 80% of its specified capacity. This allows for a long term use of the device.

Lastly the CBC050 can be combined with the CBC3150 by connecting the battery pins.

The result will be a CBC3150 with an increased storage capacity. Up to 9 CBC050 can

be added, making the total energy capacity 500 µAh. By combining several CBC3150

in this way, it is possible to charge the common battery with different inputs. This

way several sources can charge the battery, even at the same time.

All these properties make the Cymbet EnerChip ideal for this thesis and the imple-

mentation will be discussed later on.

The major downside of the EnerChip thin-film battery is the current limitation. Due

to the relatively high battery resistance (ranging from 700 Ω to 7 kΩ depending on

temperature and charge cycle, see later on), the output voltage drops significantly when

the output current is larger than 200 µA. The charge current is also limited to several

hundreds µA. Because of this, supercapacitors will be necessary to handle large input

current peaks from the solar module. External capacitors will also be needed at the

output to handle current peaks from the load. This is discussed in the section on pulse

current applications (4.3.3).

Figure 4.3 gives the battery voltage and output voltage of two CBC3150 with connec-

ted battery pins as a function of time when charging with an input voltage of 3.3 V.

The battery voltage rises quickly in the beginning but, as mentioned before, remains

almost completely between 3.8 and 4.1 V. The output voltage follows the battery vol-

tage exactly and is 0.6 V lower. This means that the output will only be 3.3 V at a

battery voltage of 3.9 V and thus the output depends on how far the battery is charged

(state of charge). This is not a problem since the rest of the circuit (see later on) can

work perfectly at supply voltages between 2.5 V and 3.6 V. Combined with the voltage

drop over the output resistance, the output voltage will eventually become too low for

the circuit to act as a supply voltage and the battery will therefore never become fully

discharged.

Figure 4.4 shows the input charge current as a function of the battery voltage. As

expected the charge current is large when the battery is almost totally discharged. As

4.3 Implementation 42

the battery is charging the charge current decreases until the battery is fully charged.

A relatively large quiescent current of 35 µA remains (caused by the enabled charge

pump) but this can be reduced to about 2 µA by pulling down the enable pin of the

CBC3150. This can be done by the digital part of the system (cf. chapter 5).

4.3.2 Supercapacitors

Introduction

All discussed energy sources apart from solar energy have a low power output. At 3.3

V and under the intented test conditions, the tested devices cannot provide currents

larger than 150 µA (At high temperatures (> 60C) thermal energy scavenging can

produce more power). Because of these low currents the cymbet thin-film batteries are

well suited as a storage device.

This is not the case for solar energy. The output power of the used solar module

(SLMD121H10 [2]) during normal daylight can be as high as 165 mW, at 3.3 V this is

a current of 50 mA. The cymbet EnerChips cannot handle these currents, the circuit

won’t be damaged but only a few percent of the input power will be stored.

Supercapacitors can be used as a storage element for solar energy scavenging. The

used supercapacitors (EDLSG155H5R5C [32] from Cornell Dubilier) have an input re-

sistance of 30 Ω, this is considerably lower than the input resistance of the EnerChips.

However, this still leads to a 1.5 V voltage drop at a 50 mA charge current, so several

of these supercapacitors need to be placed in parallel. The supplier also has supercapa-

citors available with a lower internal resistance (0.1 Ω), these supercapacitors also have

a lower maximum voltage (2.3 V rather than 5.5 V). It is characteristic of supercapa-

citors to have a low maximum voltage (around 2.3 V), the available supercapacitors

with a higher voltage rating are really two (or more) lower rated capacitors in series

in a single package. Since it are the higher voltage rated supercapacitors that have

the higher internal resistance, it can be reasoned that the higher internal resistance is

caused by the series connection of several supercapacitors. The choice was made for

the supercapacitors with the higher maximum voltage because the supply voltage of

the microcontroller is 3.3 V. With a few modifications it might be possible to use the

other supercapacitors (see later on).

4.3 Implementation 43

Figure 4.3: Battery voltage and Vout of two battery-connected CBC3150

Figure 4.4: Input charge current as a function of the battery voltage

4.3 Implementation 44

Circuit

Figure 4.5 shows the schematic of the solar energy harvesting circuit. The regulator

(LTC3105) has been discussed in section 3.1.3. It can be seen that a storage capacity

of 3 F is used (2 supercapacitors of 1.5 F) with a total internal resistance of 15 Ω.

Since the voltage across the supercapacitor will vary between 0 and 3.3 V, a boost

converter is needed to bring the output voltage of the circuit to a constant 3.3 V. This

output can then be connected to the power combination circuit (see later on). The used

boost converter is the TPS61221 [33] of Texas Instruments, this converter has an input

operating range of 0.7 V to 3.3 V. This converter is chosen for it’s low start voltage,

after an initial start-up (at 0.7 V) the converter can work from voltages as low as 0.5 V.

The circuit had problems with charging the capacitor. As soon as the supercapacitor

voltage reached 0.7V (start voltage of the regulator), the TPS61221 drawed a large

current causing the supercapacitor voltage to drop. The regulator needs a relatively

high start-up current and due to the internal resistance of the used supercapacitors,

this is not possible. This is why in figure 4.5 a 470 µF electrolytic capacitor was

placed together with a switch. The switch acts as a reset button. When the super-

capacitors are completely discharged (a case that normally shouldn’t happen since

correct dimensioning of the supercapacitor and time between transmit pulses should

prevent a complete discharge), the reset button is pressed and they become decoupled

from the LTC3105. The 470 µF capacitor is quickly charged by solar energy. When the

reset button is released, the charge on the capacitor is sufficient to start the TPS61221.

In the schematic another 2 × 470 µF capacitors are used in parallel with the super-

capacitor. These capacitors are there to handle current peaks from the load. During

high current pulses and because of the internal resistance of the supercapacitors, the

voltage at the input of the TPS61221 drops. When the supercapacitor voltage is low,

these current pulses will decrease the voltage even further. This causes the TPS61221

to go in and out of regulation which decreases the efficiency severely. The 2 × 470

µF capacitors provide the current pulses, causing the voltage to drop much less. The

TPS61221 remains in regulation, which benefits the efficiency.

Efficiency

The efficiency of the LTC3105 has already been discussed in section 3.1.3. Depending

on the supercapacitor start voltage (Vcap,start) the LTC3105 efficiency goes from 62.5

4.3 Implementation 45

% to 79.9 % (at Vcap,start = 2.0 V) and higher.

The power efficiency of the TPS61221 boost converter as a function of input voltage

(and at an output current of 95 µA) is plotted in figure 4.6. The efficiency is low at

low input voltages (40 %) and increases with increasing input voltage. The efficiency

has a maximum of 85 % at 3 V. At higher input voltages the efficiency again starts to

drop. Since the TPS61221 is a boost converter, it is not designed for input voltages

higher than the regulated output voltage (3.3V). At input voltages higher than 3.3

V, the converter still works but the efficiency is greatly reduced (< 10%, this is not

shown on the plot). The total TPS61221 energy efficiency during the discharge of the

supercapacitor from 3.3 V to 0.5 V is 71.8 %.

Just as the LTC3105 efficiency improves at a higher supercapacitor voltage (3.1.3), the

efficiency of the TPS61221 improves as well. When the total circuit is designed in such

a way that the voltage across the supercapacitor never drops below 2.0 V, the total

TPS61221 energy efficiency can be as high as 78.2 %. The downside is that this lowers

the usable storage capacity.

The total energy efficiency of both the LTC3105 and the TPS61221 strongly depends

on the minimum supercapacitor voltage. When, during daylight, the solar module

charges the supercapacitor from a voltage of 2.0 V to 3.3 V, the charge efficiency is

79.9 %. When the same capacitor is discharged during the night from 3.3 V to 2.0

V, the discharge efficiency becomes 78.2 %. This makes for a total charge/discharge

efficiency of 62.5 %. When, however, the supercapacitor is discharged to 0.5 V during

the night and has to be charged from 0.5 V during the day, the total efficiency becomes

44.8 %.

I should be noted however that with the circuit as it is now, the charge efficiency doesn’t

matter much on sunny days. The supercapacitor of 3 F is charged easily in 15 minutes

of medium sunlight (30000 lux). After the supercapacitor is charged, the rest of the

incoming energy is wasted. Therefore the total storage capacity should be increased as

not to waste energy and to increase the efficiency. But on cloudy days much less solar

energy is available and a smaller value of the supercapacitor is preferred. The smaller

capacitor charges more quickly and reaches the higher efficiency points faster. If the

intended user is satisfied with a lower storage capacity (and subsequently less frequent

data transmission), the 3 F capacitor is a good choice for every solar intensity. Wasting

residual incoming energy is not a problem if no more energy is needed.

4.3Im

plementation

46

Figure 4.5: Supercapacitors as storage element

4.3 Implementation 47

Improvements

The circuit described above is the one that has been build and tested, but there is

much room for improvement. As it is now, a supercapacitor with a maximum voltage

rating of 5.5 V is being charged only to a voltage of 3.3 V. Since the energy stored

in the capacitor is proportional to the square of the voltage across the capacitor, 2.78

times as much energy could be stored when charging to 5.5 V.

A first method to improve the circuit is to use a buck-boost converter rather than a

boost converter. The reason why this was not done in the first place is because there

are no buck-boost converters available which have a low input start voltage and because

buck-boost converters in general have a lower efficiency. The TPS61221 has a start

voltage of 0.7 V, the best buck-boost converter that was found (TPS63031) has a start

voltage of 1.8 V. It was felt that a lot of energy would needlessly have been wasted

since the supercapacitor would never have been discharged below 1.8 V. However, since

the LTC3105 has much better efficiencies at higher supercapacitor voltages and since

most energy stored in the capacitor is at higher voltages, a buck-boost converter might

be a better alternative. The LTC3105 can charge an output capacitor to a voltage of

5.25 V, the total energy stored on the capacitor will be 2.5 times higher than when the

capacitor is charged to 3.3 V.

Another possibility is, instead of using a buck-boost converter, using supercapacitors

with a lower voltage rating. As mentioned before, the supercapacitors with a higher

voltage rating also have a higher internal resistance. And the higher rated supercapaci-

tors aren’t available at higher capacitance values (several hundreds Farads). If a larger

storage capacity is needed, the choice falls on these lower rated capacitors. One large

supercapacitor could be used since the LTC3105 output is user-defined. While this

makes for a simpler circuit, it is not an ideal solution. It has already been mentioned

that both the LTC3105 efficiency and the boost-converter efficiency is low at lower

supercapacitor voltages.

It is better to place two or more capacitors in series to make a higher rated capacitor

(there are no small, high capacitance, high voltage rated supercapacitors available).

When this is done, balancing circuits are needed. This is because no 2 supercapacitors

are exactly the same and the supercapacitors in series will have a different leakage cur-

rent (at the same voltage). Because the leakage current of one capacitor will have to

flow into the other, the voltage over the capacitors will not be equally divided between

4.3 Implementation 48

Figure 4.6: TPS61221 power efficiency as a function of input voltage, at an output currentof 95 µA

them. The maximum voltage rating might be exceeded in one of the capacitors cau-

sing rapid ageing or even total failure. Linear Technology has components available to

regulate the charging of series supercapacitors but simple regulators might be designed

with a single operational amplifier. All these regulators consume extra power. This

has not been researched further.

Another improvement to the current circuit is to increase the storage capacity by in-

creasing the supercapacitor capacitance. As mentioned before the capacitors are now

fully charged in under 15 minutes direct sunlight (30000 lux). The capacitance should

be chosen so that the supercapacitor just becomes fully charged during the day, so that

no energy is wasted. However, this does not necessarily mean a large increase or even

an increase in capacitance at all. Depending on the average light intensity on the solar

module during the day, the optimal capacitance can differ a lot.

When looking at the schematic of the circuit, it can be seen that a switch is there to

decouple the supercapacitors and to act as a reset for the circuit (cf. section 4.3.2).

A better location for the switch is actually after the supercapacitor and before the

boost converter (TPS61221). A current problem is, at the moment of writing, that the

4.3 Implementation 49

start-up current of the microcontroller is too high for the 470 µF capacitor to handle.

When the switch is placed after the supercapacitors, enough energy will be stored for

the start-up.

As a last improvement, the microcontroller could measure the supercapacitor voltage

and automaticly go into low power mode when it senses that the available energy is

low. This can prevent the start-up problems caused by a completely discharged circuit.

Another possibility is to adjust the time between transmit pulses to safe energy.

4.3.3 Pulse current applications

Although the average current consumption of the microcontroller can be less than 100

µA that continuously can be supplied by the thin-film battery, high current pulses are

needed at each transmission. These currents can be as high as 25 mA (during maximum

power transmission, cf. section 5.4.6). The Cymbet EnerChip is not designed for these

currents and can only deliver about 200 µA without a significant output voltage drop

(> 0.4 V). Because of this limitation a boost capacitor has to be added at the output to

handle the current peaks of the load [34]. The value of the capacitor can be calculated

when the magnitude and duration of the current pulses are known. Figure 4.7 is a

schematic representation of the thin-film battery, boost capacitor and load during a

current pulse.

The value for the load resistance is simply the load voltage divided by the average

load pulse current. Since the microcontroller is working at 3.3 V and the average peak

current is approximately 17.5 mA, the load resistance has a value of 188 Ω. Because

the output resistance (3.2 kΩ) of the EnerChip is much larger than the load resistance

during pulses we can ignore the output resistance in the calculations for the minimum

value of C. At the start of the current pulse the capacitor is charged to at least 3.3V (Vi)

Figure 4.7: Thin-film battery, boost capacitor and load during pulses

4.3 Implementation 50

by the thin-film battery with an open-circuit voltage (Vbat) of 3.47 V (this is the open-

circuit voltage of a fully charged CBC3150). In the calculations the output voltage

vo cannot drop below 3.1 V, this is a safe value since the circuit can work at lower

voltages. As said before the average current is approximately 17.5 mA for a maximum

duration (tp) of approximately 17 ms. During discharge the output voltage is:

vo(t) = Vi · e− t

RL·C (4.1)

From this equation the value of C can be calculated:

C =tp

RL · ln( Vi

Vo,min)

= 1.45mF (4.2)

After the current pulse the capacitor has been discharged to 3.1 V and the capacitor

has to be charged by the thin-film battery to at least 3.3 V when the next current

pulse is drawn. This capacitor charge time (time between current pulses) has to be

calculated. Because there is almost no current consumption between the pulses (a

few µA), the load resistance is now high (with respect to the thin-film battery output

resistance) and can be ignored in the calculation. The equivalent circuit is shown in

figure 4.8.

The capacitor voltage is:

vc(t) = Vbat − (Vbat − Vi)e−t

Rout·C (4.3)

With Vi the initial capacitor voltage (3.1 V), Vbat the fully charged battery output

voltage (3.47 V) and Rout the battery output resistance (3.2 kΩ), the charge duration

(tbp, time between pulses) can be calculated. The capacitor is charged to 3.3 V (Vo).

Figure 4.8: Charging of the boost capacitor with a single thin-film battery

4.3 Implementation 51

tbp = RoutC · ln(Vbat − ViVbat − Vo

) = 3.6s (4.4)

These values of C and tbp are calculated for a certain current pulse profile (Ipulse =

17.5 mA, tp = 17 ms). Depending on the application this profile changes. Shorter and

smaller current pulses result in lower capacitor values and charge times. The capacitor

charge time tbp is linearly proportional to the thin-film battery output resistance. This

value changes with temperature, the value used in the equations is the measured output

resistance at room temperature. At 0°C this resistance can be 3 times higher which

results in a 3 times longer capacitor charge time tbp. Figure 4.9 shows the dependence

of the cell resistance on state of charge and temperature. The cell resistance is further

dependent on the cycle life of the battery. To lower the recharge time it is possible to

place several EnerChips in parallel.

4.3.4 Storage circuit

Power combining

Since several CBC3150 can be connected in parallel (by connecting their battery pins),

it is easy to create a common battery. This battery can be charged by any of the

input sources and acts as the main power source for the microcontroller, sensor and

transceiver. For this thesis, two CBC3150 are used, allowing for simultaneous charging

from two different sources. However, this is easily expandable.

Furthermore two CBC050 are used to further increase the capacity of the battery, re-

sulting in a total capacity of 200 µAh. Figure 4.10 gives a schematic representation of

the EnerChip circuit. The outputs of the two CBC3150 EnerChips are combined with

schottky diodes (200 mV voltage drop at a current of 40 mA) before being connected

with the microcontroller and a boost capacitor. To lower the voltage drop over the

diodes, active diodes were considered (LTC4413). Although the voltage drop and po-

wer consumption during current pulses were lower than with the schottky diodes, the

power consumption between the pulses was significantly higher. For low power pulsed

current applications the passive schottky diodes were the better choice.

It is also possible to use only the output of one of the CBC3150 EnerChips, since one

power source is sufficient. However, by combining and using both outputs, the attai-

nable output current is doubled and the boost capacitor will be charged faster. The

advantage of using only one output is that no diodes will be necessary, which eliminates

4.3 Implementation 52

Figure 4.9: Cell resistance as a function of state of charge (Cymbet application note [34])

the power dissipation of the diode.

Minimising power consumption

When the CBC3150 is enabled (Enable pin high) the internal charge pump is activated

and the battery is being charged. This is of course only the case when an input source

is available and when the battery isn’t fully charged. When this is not the case activa-

ting the charge pump will needlessly waste power and reduce the available energy for

the microcontroller. The reset pin of the device is high when the input is available and

low when it is not, therefore the reset pins of both CBC3150 devices are connected to

their respective enable pins via a large pull-up resistor.

The battery voltage (which is different from the output voltage) is read by the micro-

controller via an ADC input. The battery pin of the EnerChip is connected to the ADC

input of the microcontroller via a 10 MΩ resistor (cf. figure 4.10). This is because the

battery voltage is higher than the microcontroller’s ADC reference voltage and the 10

MΩ input resistance is used together with the external resistor as a voltage divider. To

reduce power consumption a large resistor was needed. When the battery is fully char-

ged, the microcontroller pulls down the enable pins disabling the charging. When after

a while the battery isn’t fully charged the microcontroller releases the enable pins and

it is up to the reset pins to pull up the enable pins. A hysteresis effect is implemented

so that the charge pump isn’t constantly being activated and deactivated. The charge

4.3 Implementation 53

pump will stop working when the battery voltage reaches 4.1 V and it will commence

charging when the battery voltage has dropped to 4 V. Due to the battery resistance

the measured battery voltage during charging is slightly higher than in rest, this extra

voltage drop has to be taken into account when implementing the hysteresis effect.

Efficiency

To calculate the efficiency of the EnerChip CBC3150, the device is charged with a

constant voltage (3.3 V) during 1 hour. After which the EnerChip is discharged with

a constant load. The efficiency is calculated as the ratio of the output energy over the

input energy. Figure 4.11 shows the input charge current of the circuit of figure 4.10

during a 1 hour charge period. Only one input of the circuit is used. Apart from the

beginning (first 5 minutes) the charge current drops linearly over time. This means

that charging a large battery can take a while. The charge current is limited and

even if more energy is available, the battery cannot charge any faster. Charging two

CBC3150 with 1 input source of 3.3V can take up to 2 hours, charging 4 (2 CBC3150

and 2 CBC050) can take up to 4 hours. When integrating the input current curve

(figure 4.11) and multiplying with the charge voltage (3.3 V), it can be calculated that

the total input energy is 3.13 J.

When discharging the battery, the output voltage doesn’t remain constant but it slowly

drops. The output power when discharging the battery with a constant load of 67.5 kΩ

is shown in figure 4.12. At 3.3 V this load results in a current of 50 µA, which is equal

to the average current drawn from the EnerChip when the time between transmissions

is 4 seconds. After about 100 minutes the output power (and voltage) starts to drop

faster and 20 minutes later the battery will not be able to drive the load. If the load is

not removed, the output voltage will drop to zero and go out of regulation. When this

happens the battery first has to be charged a bit before it can be used again to power

a load, even if the battery wasn’t fully discharged.

4.3Im

plementation

54

Figure 4.10: EnerChip circuit

4.3 Implementation 55

The total energy going into the 67.5 kΩ load is 1.11 J, the total energy needed for

charging the battery was 3.13 J. This is an efficiency of only 35.5%. It should be noted

however that the battery isn’t fully discharged, the battery just went out of regulation

because the discharge current became too high for the state of charge at the time.

Discharging with a lower current (and thus a higher load impedance) will allow for a

better discharge as in accordance with figure 4.2. For discharge currents as low as 1 µA

the efficiency can go up to 50 %, but this low current isn’t a realistic value for charging

the boost capacitor that follows. For currents larger than 1 µA, the efficiency is mostly

determined by how far the battery is discharged which causes the higher efficiencies at

lower discharge currents. The efficiency at currents smaller than 1 µA has not been

calculated.

When powering the microcontroller in pulse current applications, the battery will never

be completely discharged. The reason is that the EnerChip output voltage will have

dropped to far for the microcontroller to operate. This means that, apart from the

very first time, the battery will not have to be charged from a completely discharged

state. Since charging the battery becomes more efficient at a higher state of charge,

this will benefit the total charge-discharge efficiency of the battery at the expense of

the battery capacity.

4.3 Implementation 56

Figure 4.11: Input current as a function of time

Figure 4.12: Output power when discharging the thin-film batteries with a 67.5kΩ load

TRANSCEIVER 57

Chapter 5

Transceiver

5.1 Introduction

In the previous chapters, energy harvesting and storage was discussed. The purpose

of this thesis is to power a transceiver with the gathered energy. In this way a fully

autonomous module is created, which can send (and receive) data. This data contains

information about the current environment and about the module itself (e.g. battery

voltage, chip temperature). Measurements on properties of the environment are only

possible if a sensor is added.

In the next sections two different modules will be discussed. The sensor module which

will mainly broadcast its sensor data. And the base station which receives the wireless

information of all the sensor modules and sends it to the PC. The sensor module uses

energy harvesting (EH) to power itself, while the base station uses power from the USB

port of the PC. In the application (see chapter 2) of this thesis only one base station

is used, but this can be increased to enlarge the receiving range.

MicrocontrollerEnergy Harvesting

Sensor Module

SPI bus

Antenna Chip

Sensor __CS

__CS

Storage

Base Station

Antenna Chip

__CS

__CS

MicrocontrollerSPI busUART

Figure 5.1: Schematic of the complete system

5.2 Functional overview 58

5.2 Functional overview

Both modules consist of the same two major blocks (see Figure 5.1), the microcontroller

(µC) and the antenna chip. The first block, namely the microcontroller, is responsible

for retrieving analog/digital input data, digitalising it (if necessary) and processing it.

After this step, the data is sent to the antenna chip. The antenna chip will transform

the digital data into a modulated carrier frequency suitable to drive the antenna. Si-

gnals can also be received. In this case the chip will do the reverse operation and send

the data to the microcontroller.

Taking a look at the sensor module, it is clear that the microcontroller controls the

timetable (when operation starts and sensor data is retrieved, processed and sent).

This happens on a regular basis for which a timeout timer is used. In between two

timeouts, the sensor module goes to a state where it consumes little power. The µC

can also react on interrupts (button, sensor interrupt). In this case the system also

awakes from its sleep state and some data is sent. After every successful transmission

the module waits to receive an answer from the base station, which can contain settings

for the sensor module. It is possible for the base station (BS) to communicate with

different sensor modules. They each have their own SensorID which is included in the

transmitted wireless data.

In the base station, operation commences when wireless data is received. Operation

itself is still dictated by the microcontroller. When data is received, the µC gathers

this data from the antenna chip and immediately sends a (short) answer.

The communication protocol between the BS and the PC is Universal Asynchronous

Receiver and Transmitter (UART). For now this communication is unidirectional. The

data goes from the BS to the PC, but the PC does not reply. This data is processed

and displayed using a matlab Graphical User Interface (GUI).

5.3 Sensor module

This sensor module (SM) is the main purpose of the thesis. It is powered using energy

harvesting as explained in the previous chapters. This module observes a specific

property of the environment and transmits this data wirelessly.

5.3 Sensor module 59

5.3.1 Microcontroller

It is clear that the processing block should be a microcontroller. It needs to run a pro-

gram, have a low power consumption and be very small so an FPGA or small computer

are ruled out. The specifications are listed below:

• Low power: This is the most important specification because the correct opera-

tion of the sensor module relies on the amount of power that the module uses.

This power is limited by the amount that is harvested.

• Low Power Mode (LPM): To limit the power consumption even more, there has

to be a Power-Saving Mode to reduce consumption when the µC is not processing

data. This implies a low power clock which can generate an interrupt to wake up

the system.

• ADC: To measure the battery voltage an ADC is needed. The measurement of

the battery voltage does not have to be very precise (cf. section 4.3.3). As a

guideline, a resolution of 0.01 V is chosen. If the total range is 3.3 V, a minimum

resolution of 9 bits is needed.

• I/O pins: Enough pins have to be available to communicate with the antenna

chip and sensor. The ADC will need 1 or more analog input pins.

• SPI: The controller has to be able to drive the antenna chip. The most common

way to drive chips is by using SPI or I2C. SPI is easier to set-up and is faster.

I2C on the other hand uses less data lines (especially when a lot of components

communicate). Because communication is limited to only a few components, SPI

was chosen. It would be convenient if there is an SPI module on-board. Manual

implementation of SPI will be slower, less power efficient and more complicated.

• Clock frequency: There is no strict condition on speed but the data should be

processed in a reasonable time. There is a clear trade-off: a larger clock frequency

will decrease the energy intensive period, but enlarge the used power. The clock

frequency has to be compatible with the other components (antenna chip, sensor).

One of the solutions was a Cortex-M0+ Processor. The power needed stated on their

website [35] is very low (52µW/MHz), but this only includes the core, which is not a

complete controller.

5.3 Sensor module 60

A better solution is to choose a microcontroller from Texas Instruments because it is

well known for its low-power performance. Other advantages when using a component

from Texas Instruments:

• Tools: There are a lot of tools available from TI to easily program the microcon-

troller e.g. CCS[36], simpliciTI, Grace

• Online support: The datasheets are well-documented and there are lots of active

forums to discuss problems.

• Offline support: There are people in the lab who have experience with working

with TI’s MSP430 microcontrollers.

• Own experience: One of the authors already programmed some simple programs

on other devices from the MSP430 series.

• No overkill: There are other microcontrollers that can also provide correct func-

tionality, but they contain more options then necessary (and consume more po-

wer). This causes them to be more expensive and less suitable for our low cost

application.

• Availability: When buying the antenna chip (cf. section 5.4.2), an MSP430

microcontroller was included (AIR Module BoosterPack), so this means that this

type of µC will be a very good match with the transmit chip.

Eventually the MSP430G2553 was selected which satisfies the specifications (see table

5.1 and the datasheet[37]. The inexpensive, easy to use launchpad development tool[38]

was purchased, to program the MSP430. The launchpad with a plugged in microcon-

troller is shown in figure 5.2. The launchpad has a pin header to plug on an AIR

Module BoosterPack (cf. section 5.4.2). There are multiple LPM’s available. Each

of them shuts down more or less parts of the µC (CPU, clocks, DC generator). An

internal very-low-power low frequency clock is available. This clock will be used to

generate an interrupt to wake up the system.

This chip is easily programmed using Code Composer Studio[36] (CCS) which uses

C++ as language. It was also possible to use IAR VisualState[39] to program our

microcontroller, but this program is less user friendly (less support is available on the

net and in the lab). CCS is a tool from Texas Instruments, who also produced the

5.3 Sensor module 61

Figure 5.2: Launchpad Development Tool with MSP430G2553

Table 5.1: Properties of MSP430G2553

Power Consumption (Typical) (3V3)Active 230 µALPM3 with interrupts 0.5 µAADC resolution 10 bitI/O pinsDigital/Analog 8Digital 82 communication blocks possible: SPI, I2C, UARTMaster clock frequency 16 MHz (divider downto 1 MHZ)Low power clock frequency ∼12 kHz

5.3 Sensor module 62

selected chip. The tool and the chip will be attuned.

There is another tool from TI called simpliciTI[40], which enables to easily program

the MSP. This tool uses functions which are quite complicated. Some functions are

black box functions. This is not very desirable because it is necessary to know exactly

what is happening at any given moment. This makes debugging of this code very hard.

Eventually the decision was made to abandon this tool and implement the functions

from scratch. In this way, only the functions needed are programmed and the flash

memory of the microcontroller will not contain unused functions. Abandoning this tool

also facilitates debugging and gives the authors a better understanding of the processes.

The programming itself happens via UART which makes use of the internal Bootstrap

Loader (BSL) program of the µC.

5.3.2 Antenna chip

The antenna chip drives the antenna. The specifications are as follows:

• Low power: As with the microcontroller, this specification is also the most im-

portant. The chip will only be sending now and then to save power so the sleep

mode current has to be very low. The power consumption when active is less

important but should not exceed 50 mA (at 3.3 V) to limit the size of the boost

capacitors on the sensor module board (cf. section 4.3.4).

• Carrier Frequency: The carrier frequency should be situated in a frequency band

which is usable in the countries where the device will be used. To determine

this, the european table of frequency allocations[41] is used. The carrier should

also have good propagation properties. Low frequencies have better penetration

through e.g. walls. They also propagate further in the environment until a

specific attenuation is reached. On the other hand, higher carrier frequencies

typically have a higher data capacity. A carrier between 100 MHz and 1 GHz

has these required properties.

• SPI: The chip has to be compatible with the µC, so SPI communication must be

possible.

The CC110L[42] from TI satisfies the specifications. This chip is a good match with a

TI microcontroller. The chip is well-documented and online support is available. This

5.3 Sensor module 63

chip is integrated in the AIR Module Boosterpack [43] which is easily connected to

the launchpad of the MSP430 using 2 headers of 10 pins. On the boosterpack, there

are also some patches for peripheral components. The microcontroller that was chosen

is included when purchasing the boosterpack. The two components are designed and

tested to work with each other. The specifications for the antenna chip are fulfilled

with this choice (see table 5.2);

Figure 5.3: CC110L AIR Module Boos-terPack

Figure 5.4: Launchpad with Booster-pack plugged in

5.3.3 Sensor

There is a large variety of sensors that can be connected to the module to create a

useful application. A very promising application is discussed in chapter 2.

This application requires an accelerometer with free-fall detection.

Table 5.2: Properties of CC110L

Power Consumption (Typical) (3V3)RX/TX mode (not sending/sensing) ∼15 mARX/TX mode (sending/receiving data) ∼27 mASleep mode 0.2 µASensitivity ∼-115 dBm (depends on current and carrier)Maximum SPI clock 10 MHz

5.3 Sensor module 64

The specifications for the accelerometer are:

• Low Power: This is again the most important specification because only a limited

amount of power is available. The sensor has to be operational at all times, since

a free-fall situation must be detected at any given time. The power consumed by

the operational sensor has to be of the order of the power consumed by the µC

in LPM.

• Free-fall detection: It is necessary to have an interrupt which triggers when a fall

occurs. Otherwise the microcontroller should continuously retrieve data from the

sensor and process it to check for free-fall. This would require too much power.

• Compatible: Because the antenna chip and the µC communicate with SPI, it

would be convenient if the accelerometer would use the same protocol (and data

rate). The supply voltage should be the same as the other components (3.3 V),

which makes the communication and supply circuit easier to implement.

The ADXL362 from Analog Devices is the lowest power MEMS accelerometer on the

market. It performs better than its competitors from other companies such as STMi-

croelectronics, Kionix and Silica. Especially the power consumption is much lower.

The current stays well under 4 µA, even for output data rates of 400 Hz. The com-

ponent is well documented and meets the specifications. A schematic overview and

package is shown in figures 5.5 and 5.6.

The analog data from the 3-axis MEMS sensor (after demodulation and anti-aliasing)

is digitalised using an internal 12-bit ADC. This way, acceleration data is easily sent

to the µC and its 10-bit ADC does not have to be used.

The maximum absolute range is 8g (with g the standard gravity). A larger range would

be overkill and not needed for the application used in this thesis.

Free-fall detection uses the principal of inactivity detection. Inactivity means that

there is no (or less) acceleration on all the measurement axes at the same time for a

certain duration (cf. section 5.4.2). At normal static operation, the only acceleration

is gravity. If the device is in free-fall, this acceleration has no effect, so in theory

acceleration data should be zero on all axes. It is unlikely that non-zero results are

obtained due to measurement errors. So a free-fall upper bound has to be chosen. If

the acceleration drops below this threshold, the device is in free-fall.

A summary of the most important properties of the sensor is given in table 5.3. Note

that the applied supply voltage is 2V. When the sensor is plugged in the system, the

supply voltage will be 3V3, so the used power will be somewhat higher.

5.4 Implementation sensor module 65

Figure 5.5: ADXL362 Schematic overview Figure 5.6: ADXL362 Package

Table 5.3: Properties of ADXL362

Power Consumption (Typical) (2V)Operational mode, sample frequency 100 Hz 1.8 µAMotion activated wake-up mode 270 nAStandby mode 10 nAMeasurement dimensions 3 axesAdjustable range [g] ±8 downto ±1Resolution 1-4 mg/LSB (depends on the range)Maximum SPI clock 5 MHz

5.4 Implementation sensor module

The master component of our sensor module is the microcontroller, it drives the actions

of the other components (sensor and antenna chip).

5.4.1 Connections and communication

The µC receives and processes inputs from the sensor, a button, a circuit voltage level

and the antenna chip.

All inputs are digital except the voltage level. This voltage level is measured at the

battery output of the thin-film batteries. The internal 10-bit ADC is used to handle

this input. Using this measurement, good functionality of the harvesting circuit can

be ensured. When this voltage exceeds 4.1V the batteries do not have to be charged.

5.4 Implementation sensor module 66

A digital output which is connected to the thin-film batteries will disable charging.

Charging will be enabled when the voltage drops below 4V.

Another input is a send button. Whenever this button is pressed, the system will

transmit its data regardless of the regularly scheduled transmissions. This input uses

an interrupt to see if the button is pressed.

The communication between the µC, the antenna chip and the sensor is implemented

using 4-pin SPI. The master (µC) dictates the actions of the slaves (antenna chip or

sensor). The 4 pins include an active low chip select pin (CS). Data can only be

exchanged when the master puts this pin low and generates a (short) clock sequence

on the Clock pin (CLK). Data from the microcontroller to the antenna chip goes on

the Master Out Slave In line (MOSI). When data needs to go in the other direction,

the Master In Slave Out line is used (MISO). The rising edge of the clock is used to

sample the data and the falling edge to change the voltage level of the lines. Every

clock sequence is eight periods long so this corresponds with one byte of data. Further

information about SPI can be found on wikipedia [45].

The sensor communicates in the same way as the antenna chip. It uses the same SPI-

bus but a different Chip Select pin (5.1) so that it is clear to whom the microcontroller

is talking.

The configuration and selection of the pins on the µC depends on the functionality of

every pin. The communication pins of the SPI-module (CLK, MISO, MOSI) have a

fixed location. The analog input has to be connected with a pin from the P1 group This

is illustrated in figure 5.7. The specific connections are further explained in section 6.2.

5.4.2 Settings

Microcontroller

The microcontroller has an adjustable Master Clock. A lower clock frequency results

in a lower power dissipation but higher processing time. The total energy consumed

stays constant (power x time). This can be derived from a graph in the datasheet of

the MSP430G2553 [37]. This graph is shown in figure 5.8.

Lower clock frequencies are more robust against supply drops. E.g. the maximum

clock frequency (16 MHz) needs a minimum supply voltage of 3.3 V, while 6 MHz only

5.4 Implementation sensor module 67

Figure 5.7: Configuration pins of the µC, fixed SPI and analog inputs

needs 1.8 V. For the use of this thesis, the clock frequency does not have to be very

high. It suffices to take the minimum DCO frequency of 1MHz which causes maximal

robustness.

Apart from the clock system, the first major block which is enabled in the µC is the

10 bit ADC. A 2.5 V internal reference is taken, which also determines the maximum

measurable voltage. The microcontroller will manually activate the ADC and sample

the analog data. In most applications the ADC is clock sampled (automatic sampling

every x µs). This is not be very useful because only when data is sent, should the

ADC determine the voltage at the input. An extra timer would also mean more power

consumption.

The second enabled block is the SPI block. This block provides communication with

the antenna chip. The SPI bus is clocked at the maximum available frequency (1 MHz).

The bitrate is chosen to be 100 kbps.

Antenna chip

Every time the sensor module starts up, the µC uses SPI to change the default values

of the registers of the antenna chip. These registers determine the operation of the

chip. Most settings are found using SmartRF Studio [46]. The choice and meaning of

the most important settings are explained below:

5.4 Implementation sensor module 68

Figure 5.8: Active Mode Current vs DCO Frequency

• Transmit power: This property determines the functional wireless range. Prac-

tical measurements are given in table 5.5 of section 5.7.2. The perimeter of the

range is not a clear line but consists of a grey zone where coverage is not gua-

ranteed but most of the packages arrive correctly. The start and end of this zone

is shown. At the end of the grey zone, there is little or no coverage. Indoor

measurements depend mainly on the type of building and specific ground plan.

The maximum power that is allowed is determined by the FCC (Federal Commu-

nications Commission). A summary of these rules for our carrier frequency (see

next setting) is given in [47]. Following these rules the maximum transmittable

power is 30 dBm since the maximum antenna gain is 0 dBi [48].

• Carrier Frequency: There are three possible frequency bands which can be used.

The first band reaches from 300 to 348 MHz. According to European ECC

(Electronic Communications Committee) [41] most of this band is intented for

air control (defence and civil) and radio astronomy. This band is not suitable for

the system. The second band (387 - 464 MHz) has more opportunities. There is

room for Personal Mobile Radio (PMR), amateur radio and there is an ISM-band

5.4 Implementation sensor module 69

(Industrial Scientific and Medical) (433.05 - 434.79 MHz). In the last band (779

- 928 MHz) there is some room for alarm systems. The best option is the second

band and more specific the ISM band. Transmitting in this band is free of charge.

Note that in the US the 915 MHz ISM band can also be used.

• Modulation format: GFSK (Gaussian Frequency-shift keying) is used. This code

does not suffer from amplitude fading in contrast with ASK (Amplitude-shift

Keying). GFSK has a higher spectral efficiency than FSK due to (Gaussian)

pulse shaping. This shaping removes the sharp transitions from FSK. The relative

distance between 2 symbols is larger than 4-FSK which improves the BER.

• Data rate: The data rate does not have to be very high. At 250 kBaud, the

data is frequently bad received. A very low data rate (2.4 kbaud) causes a long

unwanted delay (3,33 ms/byte) during high power consumption. There is a clear

trade-off between robustness (low data rate, which causes higher sensitivity: table

7 in the CC110L datasheet [42]) and power consumption (high data rate, small

delays). A data rate of 100 kBaud is used.

• Sensitivity: It is possible to reduce power consumption, if the sensitivity of the

sensor module is reduced when sensing and receiving. This reduced sensitivity

will not decrease the operational range of the sensor module because the BS will

be sending at its maximum power +10 dBm since the BS does not depend on

energy harvesting. It is connected to a computer. The chosen data rate results

in a sensitivity between -95 dBm and -104 dBm according to table 7 from the

datasheet[42].

Sensor

At start-up the microcontroller also sets the registers of the sensor to enable free-

fall detection, and start acceleration measurement. The most important settings are

explained below:

• Inactivity threshold: Free-fall detection uses this threshold. When the accelera-

tion on all axes is lower than this threshold for a certain amount of time (see

inactivity time), an interrupt is generated. For now this value is set to the large

value of 600 mg to enable easy debugging but this has to be optimised for real

life scenarios.

5.4 Implementation sensor module 70

• Inactivity time: This register is also used by free-fall detection. This value states

the time (number of samples) for which the acceleration should be below the

inactivity threshold. This value is set to 30 ms, but has to be optimised for real

life events.

• Measurement range: The range can be set to ±2g, ±4g or ±8g. For the appli-

cation that is chosen, the three options have enough resolution to perform well.

For now the range is set to the maximum (±8g) but this can be changed. This

can be useful when e.g. data has to be read and processed using an algorithm to

determine if the fall is legitimate (see 5.9).

• Output data rate: This determines the sample rate at which the data is placed

into the FIFO buffers. This rate is also determined by the conditions the data

should have for a potential algorithm (legitimate fall).

Energy optimisation

After a functionally correct system was constructed, changes were made to make the

design less power consuming. The most important changes are listed below:

• Burst sending: It would be very energy inefficient to keep the µC and antenna

chip running at all times. When the system is inactive, the antenna chip should

be disabled and the µC is set to a Low Power Mode. There are 4 LPMs possible.

LPM4 is the least consuming because all clocks and the CPU are disabled and

the system only reacts on interrupts. This mode is not suitable because for the

purpose of thesis, the system has to send data on regular basis. A better solution

is LPM3. In this mode, a very low power clock can be active. It will drive Timer

A. Using this timer, it is possible to generate a regular timeout interrupt that

awakes the system. The low frequency auxilary clock (ACLK) is used (12 kHz).

When going into LPM3, a register counts the clock cycles and triggers when a

preset value is reached. In case this value is larger than the register capacity,

multiple shorter timeouts are used to reach the desired longer timeout value.

The system can also be awakened using a sensor interrupt (cf. section 5.3.3).

A third way to awake the system is by using the interrupt of the button. After

these forced interrupts, the timeout timer is reset (timeout will occur again after

the preset value of before).

• Sampling rate: The time in between two bursts is determined by the available

power. When data is sent (received) over the antenna, the amount of immediately

5.4 Implementation sensor module 71

needed power is larger than the amount of power being harvested. So in between

2 samples, enough power has to be stored (in boost capacitors) to absorb this

sudden burst. After every burst, the boost capacitors are (slowly) recharged to

3.3V using the thin-film batteries.

• Delays: To generate delays, the intrinsic function ( delay cycles()) is not always

used. This would consume more energy than using a clock with an interrupt.

The low power auxilary clock (12 kHz) is mapped on Timer A. For large delays

(see above: Burst sending), LPM3 can be used.

• Ports: When a port of the microcontroller is not used, it will be placed to the

output direction and pulled to GND. This avoids extra power consumption.

• Register writing: Normally the registers were written sequentially. Every register

has its own write operation: CS goes LOW, write the address, write the register

data, CS goes HIGH. This can be done much faster and efficient using burst

register access. This means that all the registers with consecutive addresses are

written in one write operation. After CS goes low the first adress is written with

a burst bit high. Now the register data is written one by one. The write address

is incremented internally in the antenna chip. If a lot of registers are written this

way, the total time (power) is halved.

5.4.3 Coding

A schematic overview of the implementation is displayed in figure 5.9. The different

blocks are discussed below.

Initialisation This block will properly set the settings of the microcontroller and the

registers of both the accelerometer and the antenna chip.

The settings for the µC are determined using Grace[49]. This is a pre-installed plugin

in the newest versions of CCS (5.1 and later). It is possible to graphically configure the

options, modules and I/O pins of the microcontroller. When a build is commenced,

CCS will create a main Grace file with multiple subfiles in which the correct values

are written to the correct registers. After this, the next lines of code initialise the

accelerometer. The registers will be written using the SPI-bus.

The registers of the antenna chip are set in a similar way in the next step.

Now the environment is set and interrupts can be initialised and enabled.

A SensorID is created. The four least significant bits are fairly random generated using

the last bits of the ADC. The four most significant bits are hard coded. The chance

5.4 Implementation sensor module 72

Initialisation

Operation

Read and ActionsFlush buffers

LPM

Wake up

Interrupt

Flush buffers

Set registers Antenna Chip CC110L

Set registers Accelerometer ADXL362

Configure the MSP430 settings (Grace)

µC to LPM3

Reset interrupts

Control system (ADXL362 and Batteries)

Disable µC modules (SPI, ADC10)

Control system (ADXL362 and Batteries)

Disable/Sleep CC110L

Abort LPM3

Enable µC modules (SPI, ADC10)

Enable/Wake CC110L

Wait for Answer (~10ms)

Send data (battery level, interrupt)

Check thin-film batteries

Generate SensorID

Initialise and enable interrupts

YESNO

Figure 5.9: Schematic overview code for the sensor module

5.4 Implementation sensor module 73

of two sensor modules having the same sensorID is 7%. If the IDs are equal, reset

one of the SMs and another ID will be created. This code is a temporally measure to

facilitate programming (the same code can be used for the two sensor modules). In

real life situations (more sensor modules) the sensorIDs will hard coded differently for

every SM or a larger random ID is generated. Using the sensorIDs, the BS is able to

distinguish between the different sensor modules (cf. sections 5.5 and 5.8).

Operation This block contains the processing part of the program. Before the code

commences, a while loop is started which includes all the following steps.

First the analog input (from the thin-film batteries) is checked, whether or not the

value exceeds the limit to which the batteries need charging. Whenever the battery

voltage exceeds 4.1 V, charging is disabled. When this voltage drops below 4 V charging

should be enabled. This enabling/disabling happens in the LPM block.

In the next step data is sent to the FIFO buffer of the antenna chip and a transmit

signal is given to this chip. This data contains the SensorID and information about

button pushes, free-fall detection and the measured voltage level from the previous

step. Other information can be added, e.g. data about the acceleration history around

a free-fall event, temperature.

After every transmission, the microcontroller will set the antenna (chip) to receive

mode for about 10 ms. In this time an answer from the BS can be received.

When an answer is received, the data will be read from the FIFO buffer of the antenna

chip and then processed e.g. changing transmit power or transmission interval. If not

(or an erroneous answer) the FIFO buffers are flushed.

LPM In this part of the code the different components of the module are disabled

or set to a Low Power Mode.

The first component is the antenna chip. A power down command is given. The power

down state will stay active until the chip select pin goes low.

In the next lines of code, the SPI and ADC modules are disabled.

The next step is to make sure the accelerometer is in a correct state, so that it can

generate an interrupt whenever a free-fall is detected. Another action in this step

is enabling/disabling of the charging of the batteries according to the result of the

operation block. After this, interrupts are enabled in the microcontroller. It will use

these interrupts to awake from its LPM.

In the last step of the LPM block, the microcontroller puts itself in sleep mode. LPM3

is used (cf. section 5.4.2).

5.4 Implementation sensor module 74

Wake up This block will wake the system after an interrupt was observed. Note

that the microcontroller can still detect interrupts while it is in the sleep state. The

first part of code will wake the microcontroller. This happens inside the corresponding

interrupt routine (fall, button, timer). Less than 1 µs is needed (see datasheet[37]).

The modules which were disabled in the previous block are rebooted. Grace recom-

mends a long delay (30 ms) to ensure that the reference voltage is stable.

In the next line of code, an SPI command is sent to awaken the antenna chip. The

specific command does not matter because, the wake up mode is triggered on the tran-

sition of the chip select line. After this step, the system goes back to the operation

block and starts again.

5.4.4 Coding and testing issues

There were a lot of small problems when coding and testing the functionality which

increased the development time of the code. Some of them are explained below.

• RFstudio problems: A certain register was set to enable extra communication

with the microcontroller. This pin was set to a low output on the microcontroller

(to reduce power consumption). Whenever the antenna chip wanted to send

something on this pin it caused a short, which resetted the system. The values of

the registers are noted in a mix of hexadecimal and decimal values without any

indication.

• CCS Debug: Debugging of the code using the debug function of CCS can some-

times lead to very strange situations when the code is stopped. Eventually the

debug functionality was abandoned and errors were visualised using the digital

oscilloscope.

• Calibration: Now and then, the antenna chip has to calibrate its oscillator. Du-

ring calibration, no commands can be sent to the chip. In the first versions of

the code, this behavior was not taken into account. This caused malfunctioning

of the antenna chip. The solution for this problem is repeatedly polling to the

antenna chip to check its (calibration) status until calibration is over.

5.4.5 Board

For testing the transceiver separately (without Energy Harvesting and Storage) a de-

velopment board was used, namely the AIR Module Boosterpack (cf. section 5.4.2).

It is possible to plug the microcontroller onto this board which gives us a small and

5.4 Implementation sensor module 75

testable PCB. The available patches were used to solder pull up resistors, decoupling

capacitors, a button and a sensing wire. This board is shown in figures 5.10 and 5.11.

Figure 5.10: Simple board sensor mo-dule with accelerometer

Figure 5.11: Simple board sensor mo-dule without accelerometer

5.4.6 Power performance

When the transceiver is in low power mode the power consumption is constant. The

system is powered by a 3.3 V supply and the current is 2.7 µA . This results in a power

of about 9 µW. Most of this current is consumed by the active accelerometer.

When data has to be sent, the current increases to a couple of mA and follows a cer-

tain profile (a burst) depending on the presence of a base station within reach. These

profiles (transmit power 0 dBm) are displayed in figures 5.12 and 5.13. Note that the

current that runs into the microcontroller is negligible compared to the peak currents

caused by the antenna chip.

The two profiles are quite similar but they differ in the fact that in the first one an

answer was received. They both have the same start-up peak. This current is used

to wake up the different components and enable the internal oscillators. After this

peak, a smaller current is consumed (about 1.7 mA). This is caused by the internal

voltage regulator and the crystal oscillator of the antenna chip. After a while (3 ms)

the system is ready to send its data. The send peak consist of 3 stages. The first stage

is a current level which corresponds with the FSTXON state (a transitional state). In

this stage the frequency synthesiser is running and calibration can be performed (8.4

mA). The next stage (about 16 mA) is again a transitional state in which the carrier is

prepared to drive the antenna. The actual sending happens in the third stage, where

a maximum of 26 mA is consumed.

After this, the microcontroller commands the antenna chip to listen to an answer.

5.5 Base station 76

Whenever a BS is in range, there are again 3 stages in the receive profile. The first

one is (analog to the first send stage) a calibration stage (8.4 mA). In the second stage

the antenna is sensing for an answer for a maximum of 10 ms at a current of about

16 mA. This waiting time is a set in the microcontroller. If an answer is received the

current increases to 25 mA for a short time and then decreases to 1.7 mA. This current

stays constant until the listening time (10 ms) is over. After this, the antenna chip

shuts down. If no BS is present the full 10 ms of listening time is used to sense and

the current remains constant at 16 mA.

The energy used when sending and receiving can be calculated by integration of the

current (formula 5.1).

E =

∫P · dt = 3.3V

∫I · dt (5.1)

This integration is performed discretely. When the BS is in range, the consumed energy

of one burst is 0.5 mJ. When no answer is received this energy is 0.9 mJ (due to longer

sensing). This energy burst has to be delivered by the boost capacitors (see 4.3.3) at

the supply voltage in order to prevent an excessive voltage drop of the supply.

5.5 Base station

The second module is the base station. This module is designed to receive the data

from the sensor modules. After every reception a short answer is sent back to the

sensor module. This device is not power limited since it is powered using the computer

via the USB port so no special low power modifications have to be made.

5.5.1 Components

The module consists of the same two important components as the sensor module: the

µC and the antenna chip (cf. figure 5.1). This facilitates the interchangeability of the

components of the BS and the sensor module.

5.6 Implementation BS

The microcontroller dictates the actions of the antenna chip and communicates via

UART and USB with the computer.

5.6 Implementation BS 77

Figure 5.12: Current profile sensor module, BS in range, 0 dBm

Figure 5.13: Current profile sensor module, no BS in range, 0 dBm

5.6 Implementation BS 78

5.6.1 Connections and communication

The only inputs the µC of the BS has, are the ones provided by the antenna chip and

a button press. There are no sensors attached and no voltages have to be measured.

The microcontroller and the antenna chip communicate in the same way as in the sen-

sor module (SPI).

The button can be used to alter the answer towards the sensor module after a successful

reception. This can imply e.g. a change in transmit frequency or power from the sensor

modules. There is also a reset button available on the launchpad which grounds the

RST (figure 5.7) pin if pushed. The microcontroller will reset and start at initialisation

(cf. section 5.6.3).

The communication between the module and the PC happens using the UART module

which is also on-board the µC. The microcontroller stays plugged into the Launchpad.

The USB-interface from this launchpad provides a 9600-Baud serial UART connection

with the PC. For the use of this thesis, the microcontroller only sends data towards

the computer because it only has a monitoring function.

5.6.2 Settings

The microcontroller settings are not the same as in the sensor module.

E.g. in this module the 10 bit ADC is not enabled, because no voltages have to be

measured.

The clock is set to 1 MHz. This allows a fast enough processing speed. If at a later stage,

more processing has to be done (algorithms), the clock frequency can be increased. This

can be done freely because now a stable (no drops) supply voltage is available (PC).

The SPI block is also active and communicates with the antenna chip. The frequency

at which it works, is the frequency of the master clock of the microcontroller. For

faster communication, this clock can be increased to a maximum of 10 MHz. This is

of course only possible if the highest clock frequency of the microcontroller is changed.

The low power low frequency clock (12kHz) is not active because no long delays are

needed and enough power is available to use the intrinsic function delay cycles().

The UART block is enabled to communicate with the computer. The baud rate is

chosen to be the maximum 9600 Baud. It would be less time consuming if it were

5.6 Implementation BS 79

higher but the rate is limited by the capacity of the USB interface of the launchpad.

A practical solution for this waste of time can be found in section 5.6.3.

The registers of the antenna chip are set to the same values as in the previous module.

There are however some small differences. The most important ones are:

• Transmit Power: Power consumption is not the limiting factor in this device so

a higher output power is chosen (+10dBm).

• Sensitivity: The reduced sensitivity setting is not used which will give a +2dB

sensitivity increase compared to the sensor module.

5.6.3 Coding

A schematic overview of the code for the microcontroller is given in figure 5.14. The

code now consists of two major blocks.

Initialisation This first block is only executed once at system start-up. The first

step is to initialise the MSP430. Grace facilitates the code for configuration of the clock

module, the I/O pins and the SPI module. The next step is to set the registers of the

antenna chip. The SPI communication channel is used to send the write commands.

The registers are determined using SmartRF Studio [46].

Now that the system is set, interrupts can be enabled. There is only one interrupt,

which is the button.

Operation The second block is started with the initialisation of a while loop. This

while loop causes a repetition of the operation block. The first step of the block is a

check whether or not data is received. If not, the system does nothing and skips the

four following steps. If data is received, which can be detected by a change in status

of the antenna chip, the data is read from the FIFO buffer using the SPI interface. In

the initial code the data which was received, is immediately sent to the computer using

UART. After UART communication ends, an answer is sent to the sensor module. Ho-

wever, because the UART communication takes a lot of time, the last two steps were

switched.

After the received data is stored on the microcontroller, the answer for the sensor

module is sent to the antenna chip and transmit mode is enabled. While the chip is

5.7 Performance 80

transmitting, the UART communication is executed. This results in a parallel UART/-

transmitting state which decreases the time that the BS is unable to receive data from

other sensor modules.

The UART communication (only 4 bytes, equals about 4.5 ms) takes less time than

sending the wireless data (about 5 ms), so when the UART communication is complete,

the microcontroller repeatedly checks the status of the antenna chip until it is IDLE

again (for about 0.5 ms). This event means that the transmission of the wireless data

is complete.

The last step is performed in every loop. This step checks if an erroneous byte was

received. If so, the buffers of the antenna chip are flushed and the system is reset to

receive mode.

A delay is added at the end which determines the polling frequency.

5.7 Performance

5.7.1 Communication

Implemented communication algorithm

For now, the communication algorithm between the base station and the sensor modules

is quite basic. It is only half duplex so two modules (BS or SM) cannot send at the

same time.

During normal operation of the system, the BS will be in receive mode. There are only

specific moments when the BS will send. Every time it receives data from a sensor

module, the BS is able to respond with a short answer which includes some settings

for the corresponding sensor node.

During normal operation of the system, the sensor nodes are in low power mode. While

in LPM, they cannot receive or send data. Every node will awake when its preset timer

ends or when a forced interrupt is triggered (button or fall sensor). The node sends

its data message and waits a couple of milliseconds for an answer. When an answer is

received, it will be processed. Eventually the node will go to LPM again.

In figures 5.15 and 5.16 this communication is visualised by monitoring the data ac-

tivity between the components of the different modules. One communication burst is

displayed. The allocations of the data lines are displayed in table 5.4.

5.7 Performance 81

Initialisation

Operation

Flush buffers

Set registers Antenna Chip CC110L

Configure the MSP430 settings (Grace)

Delay

Check if data has been received

Initialise and enable button interrupt

NO YES

Do nothing Read data from FIFO

Send data to FIFO

Send UART

Wait for IDLE

Flush buffersCheck for error buffer byte

Figure 5.14: Schematic overview of the code for the base station

In the first part of the communication (figure 5.15) the sensor module wakes up and

puts its data in the FIFO buffer of the antenna chip (SPI). This data is then transmit-

ted by giving the correct SPI command. The microcontroller then checks repeatedly if

the antenna chip is still sending. If the transmission is completed, the antenna chip is

set to receive mode and the microcontroller waits for 10 ms.

When the data is received by the antenna chip of the base station, the next poll will

trigger the microcontroller. The µC requests the data from the FIFO buffer from the

antenna chip (SPI). Next, new data will be send to the antenna chip and transmit

mode will be activated. While the antenna is transmitting, the received data (from the

5.7 Performance 82

Table 5.4: Allocation of the data lines

Digital probe Data line

D0 Sensor module: SPI SOMID1 Sensor module: SPI SIMOD2 Sensor module: SPI CLK

D3 Sensor module: SPI CSD8 Base station: UART TXD9 Base station: UART RXD10 Base station: SPI SOMID11 Base station: SPI SIMOD12 Base station: SPI CLK

D13 Base station: SPI CS

sensor module) will be send to the PC using UART. Once this is done, the microcon-

troller will check if the wireless transmission has been completed.

In the second part (figure 5.16) the transmission at the base station is over. The BS

goes to receive mode again and starts polling again for new received data.

At the sensor module side, the data from the BS is received. After the predefined

delay (10 ms), the microcontroller checks whether or not the antenna chip contains

valid data in its FIFO buffer. This data is read and processed. In the next step, the

sensor module goes to sleep mode.

Possible communication conflicts

The base station cannot send while the modules are sending. Any data from a sensor

module while the BS is transmitting will not be received and will be lost. The sensor

to which the answer from the BS was intended will get either erroneous data, the data

from the sensor module or the correct answer from the BS. Which one depends on the

distance between the components.

Another problem that can occur is when two sensor modules send at the same time,

the data of at least one (the weakest) will be lost.

Using another, more sophisticated communication algorithm will solve those problems

(cf. section 5.9).

5.7Perform

ance83

Figure 5.15: Part 1: SPI and UART communication of SM (D0-D3) and BS (D8-D13)

Figure 5.16: Part 2: SPI and UART communication of SM (D0-D3) and BS (D8-D13)

5.8 Matlab 84

5.7.2 Power

The transmit power of the sensor module determines the reception range. The base

station replies with full transmit power, so the sensor module does not have any pro-

blems with receiving the answer. In table 5.5, some possible transmit powers are listed

with their corresponding ranges. The range is characterised by two lengths, the start

and end of the grey zone (cf. section 5.4.2). Note that these measurements were per-

formed outdoor with some reflection of the ground and surroundings.

It can be concluded that a power of 0 dBm has a large enough range for the applica-

tion in mind. Nevertheless, this range will decrease drastically when measured indoor.

Some measurements were performed indoor. At 0 dBm, the signal was received across

multiple rooms.

5.8 Matlab

The base station sends its data to the computer via UART. This data is received and

processed using Matlab. A GUI is used to visualise the received data.

On start-up of the GUI, the variables are initialised and the interface of figure 5.17 is

shown. First of all the correct COM port has to be set at which the BS is connected.

When the START/STOP button is pressed a while loop commences that opens a serial

communication at the specified port. Then the code will repeatedly check if a byte was

received and if this byte is equal to 0. The data from the BS always starts with a zero

byte followed by a SensorID that differs for every sensor module. This way, Matlab

can differentiate between the different modules. After this overhead, the real data is

retrieved and analysed. For now, there are only 2 bytes of data. The first bit represents

the presence of a button press. The second bit determines whether a fall has occurred.

Then there are 4 unused bits. The last 10 bits are needed to represent the voltage level

of the thin-film batteries. The voltage levels of the batteries from the sensor modules

are plotted in the graph. If a fall or button press occurs, it will be indicated on the

graph (respectively a star or a circle). The serial communication is closed whenever

the START/STOP button is pressed again. Reset will restart the GUI and erase all

the data.

5.9 Improvements

In this section, some of the features are discussed which couldn’t be, or were not yet,

implemented as well as some additional possibilities of the modules. The features which

5.9 Improvements 85

Table 5.5: Transmit power versus receive range at 433 MHz in an outdoor situation

Transmit Power [dBm] Start grey zone [m] End grey zone [m]

-30 7 13-20 20 27-15 32 50-10 50 70-6 65 1000 110 150

Figure 5.17: GUI

5.9 Improvements 86

were not optimised entirely are caused by the limited time frame of this thesis. The

transceiver is only a part of the entire system. Debugging the system also costed a lot

of time e.g. getting SPI to work using simpliciTI (which was abandoned) and creating

an initial simple communication link. Below, some of the possible improvements are

listed:

• Adapt Power: It is possible to adapt the send power of the sensor module by

writing to the Power Table register. If no answer is received from any base

station, the sensor module can opt to increase its own transmit power on the

next wake-up to try to reach a more distant BS. If a succession of successful

interactions between the BS and the SM occur, one can opt to try and reduce

the transmit power of the SM to reduce power consumption.

• Fall algorithms: There is an option in the accelerometer to request acceleration

data around a certain interrupt. In this case a fall detection. Using this data,

it can be decided whether or not the fall was legitimate. The question whether

these algorithms should be performed on the sensor module, on the BS or on the

PC, will depend on how long the processing takes and how much data is needed

to determine a fall. If a lot of data is needed, the sending of it will be very power

consuming so then it would be best to do the processing in the sensor module.

If not, the data can be easily send to the PC/BS. If a long processing time is

needed, the sensor module will be online for a longer period which again increases

power consumption.

• Faster UART: Now, the maximum baud rate at which communication is possible

is 9600 Baud. This is limited by the Launchpad on which the BS is connected.

If a custom made UART/USB interface is made this rate can be increased to

the maximum baud rate of the computer/Matlab. This cannot be higher than 2

MBaud, which is the limit of the microcontroller. This 2MBaud is only possible

if the clock frequency of microcontroller is increased.

• BS settings: For now, the BS has very similar settings as the sensor module

(some exceptions exist). The BS can be optimised for better performance e.g.

the clock frequency can be set to a higher value because a steady supply voltage

is available, the SPI clock frequency can be changed. This can also be done in

the sensor module.

• Sensor settings: The settings of the fall detection of the accelerometer are set

to generate an interrupt quite quickly (cf. section 5.4.2). This is convenient for

5.9 Improvements 87

easy debugging. For a real life scenario, these settings have to be altered and

optimised.

• Include sensorID: If a sensor module has sent its data, it will listen to any response

the BS sends back. If multiple sensor modules are present and listen at the same

time, they will both receive the same message and interpret it as the answer to

them. This can be avoided if the sensorID is included in the answer.

• Module: Texas Instruments also has another module which would be suitable for

our use which is the EZ430RF2500 [50]. This module is a premade, small, all in

one module which sends on a higher frequency (about 2.4 GHz). The prices are

equivalent but a higher frequency means less penetration through walls which

makes it less suitable for indoor use. A first advantage of this module is that,

in a later state, the system might be integrated into the network card of the

computer and communicate using Wifi frequencies. Higher frequencies typically

have a higher data capacity so this would also benefit the communication. This

has to be examined further.

• Frequencies: It is possible to change the frequency of the sensor module quite

easily. The BS can command every sensor module to use another frequency to

cancel out interference. Another possibility is to use a different frequency for the

uplink and downlink. Now no interference can happen between BS and sensor

module. In any case multiple BSs are needed with each a different task/fre-

quency. This can also be implemented using one BS with multiple antenna chips.

SmartRFstudio suggests a channel spacing of about 0.2 MHz. This means that

in the 433 MHz ISM-band (bandwidth 1.74 MHz) eight channels can be used.

• Duplex UART: For now the PC works as a monitoring station. If it is desired

that the PC should be a control centre the UART communication should be

duplex. If it is possible to send data to the BS, the BS can alter the settings of

the different sensor modules by sending them a different answer.

• Receive interrupt: Both the BS and the sensor module do not use a receive

interrupt. The microcontroller of the BS repeatedly polls the antenna chip if

data was received. The sensor module just waits for 10 ms, and then checks if

data was received. If an interrupt from the antenna chip is used, some time can

be spared.

• Free-fall: If a free-fall occurs, the sensor module will transmit this data. If no

answer is received, the sensor module goes to LPM and does not remember the

5.9 Improvements 88

free-fall event. This is easily solved by remembering this event and sending the

message multiple times until it is acknowledged. The timeouts between trans-

missions should be temporally decreased to get a warning to the BS as soon as

possible.

• Communication algorithm: A more sophisticated protocol would solve the com-

munication conflicts. There are standard protocols available for wireless networks

but most of them do not account for low power applications. It is out of the ques-

tion to keep the sensor modules listening for an extended period of time. A more

realistic approach is given below.

The protocol consists of a sort of time division multiplexing. An example of the

timeline of the protocol is displayed in figure 5.18. The BS determines the time

schedule when the sensor nodes are allowed to send. To insure communication

is over when every interval ends, some margin is added. In each communication

burst to a specific sensor node, the schedule can be made clear to this node.

Internally the sensor module knows in what interval it is allowed to send using

its own timer. The sensor module does not have to send something in every, for

him scheduled, interval. Due to the time division, the (scheduled) sensor modules

cannot interfere.

Sending on interrupt (button or fall) will be handled during the ‘free’ intervals

which is also divided into multiple slots (not shown in figure 5.18). ‘Free’ means

that it are intervals in which every sensor module is allowed to send. In these

intervals interference can occur and when it does, the sensor modules will not

get a valid answer. They will resend their message again in the next ‘free’ in-

terval (maybe this time in another randomly chosen slot) or in the for them

scheduled interval (whichever is earliest). On a scheduled interval, interference is

nonexistent so then the message will definitely be received and answered (if the

BS is in range).

There is however a problem with new sensor modules that will try to connect

with the BS. If it is possible (enough power is available) the sensor will listen for

an extended period of time to intercept any messages, figure out the schedule of

the BS and locate the ‘free’ intervals. In those intervals the new sensor module

can communicate with the BS an can so be included into the schedule. If not

5.9 Improvements 89

enough power is available to initially listen for an extended period of time, the

sensor module will transmit at periodic time intervals. Whenever this sending

happens in a ‘free’ interval, the BS can reply and include the sensor module

into the schedule. Remark that when a new sensor sends data during scheduled

intervals of another sensor module, the messages will interfere. This has to be

detected by the corresponding scheduled sensor module and it will send its data

again in a later interval. Note that the interval at which the new sensor module

initially sends has to be chosen correctly to avoid interfering with the same sensor

module over and over again.

If a sensor module does not receive a valid answer after several transmissions (es-

pecially during its own scheduled interval),the sensor module can opt to increase

its transmit power. This failed transmission can be caused by an interfering sen-

sor module or being out of the receiving range of the BS. The last problem can

be solved by increasing the transmit power of the sensor module. The problem

of interference should solve itself in time (due to choosing random slots in the

‘free’ intervals) if the sensor network is not too crowded.

Sensor 1S & R

SEND & RECEIVE = S & RMARGIN = M

Sensor 2S & R

Sensor NS & RM M M

Sensor 3S & R 'FREE' Interval Sensor N+1

S & R

Sensor MS & R

Sensor 1S & RM M

Sensor 2S & R M

M

Time

'FREE' Interval

Figure 5.18: Protocol timeline

TOTAL SYSTEM 90

Chapter 6

Total system

6.1 Introduction

In this chapter the developments and research presented in the previous chapters are

combined to create one sensor module on a PCB. Some of the problems which occured,

will be discussed as well as some of the design choices.

6.2 Boards

Three new boards are created (cf. figures in section A.1 of the appendix) . The main

board (sensor module board) contains four thin-film batteries, the boost capacitors and

the digital part (microcontroller, accelerometer and antenna chip). The other boards

each contain a system to harvest energy from a different source (thermal and solar). To

these boards, the harvesting elements can be connected (Peltier element, solar module).

The boards themselves contain a regulator and supercapacitors. The step-up converter

is also present in both boards, but on the thermal board, this converter is included in

the regulator chip. The thermal and solar board can be easily plugged in below the

sensor module board (cf. 6.1 and 6.2) This reduced the size of the total sensor module.

Some extra measurement pins (two analog and two digital pins) are transferred from

the EH boards to the SM board. For now, the microcontroller does not read those

values. A reset button is added which resets the microcontroller.

The antenna chip is set in a corner, with the antenna facing away from the rest of the

circuit. Therefore the amount of disturbance of the radiation pattern of the antenna,

due to the surrounding circuit, is limited.

6.3 Performance and measurements 91

Figure 6.1: Complete sensor module(top view)

Figure 6.2: Complete sensor module(side view)

Solar cell Regulator

Super-Capacitor

Step-up converter

Thin-film batteries

Boost-Capacitor

Microcontroller

Peltier element Regulator

Super-Capacitor

Sensor

Antenna chip

Thermal board

Solar board

Sensor module board

Figure 6.3: Schematic overview of the sensor module and its supply modules

6.3 Performance and measurements

If the energy harvesting elements are connected, the system charges correctly. If no

solar or thermal energy is available, the thin-film batteries are charged using the remai-

ning energy on the supercapacitors. Meanwhile the microcontroller stays operational

at all times.

When the sensor module falls, multiple interrupts are sent to the microcontroller (due

to the low fall threshold). The microcontroller will send multiple times in less than

a second. Due to the dimensioning of the boost capacitors, this will not cause the

voltage to drop below the minimum supply voltage of the digital part of the system.

If the supply voltage drops below 2.5 V, there will be consequences concerning the

measurements of the ADC which needs an internal 2.5 V reference voltage.

6.3 Performance and measurements 92

To check how long the system lasts, and determine the transmit timeout of the micro-

controller, the microcontroller is set to LPM. Only the solar board was connected to

the SM board. At the start the supercapacitors are charged to 3.5 V (not maximal)

and the thin-film batteries to 4.07 V (maximal). This state is achieved when the sen-

sor module can charge for 45 min in full sunlight. Now the solar cell is disconnected

and the system starts to discharge slowly. After more than a day, the supercapaci-

tors do not contain any useful charge (about 0.5 V). The thin-film batteries are still

fully charged. Now the thin-film batteries power the digital part of the system. After

another day the thin-film batteries cannot supply the system any more. So, if no EH

is available and the storage is full, the system can stay operational (an interrupt will

still trigger a transmission) for one extra day if the system is in LPM. This result is

very convenient for our application. If no EH is available and the system does not have

to send frequently (e.g. at night) the system can go to LPM and last for at least 2 days .

When the supercapacitors are charged to 3.5 V and the thin-film batteries are full,

the system can last for 12 hours if the sensor module transmits every 6 seconds. This

is long enough to last during the night, but the supercapacitors will be totally depleted.

Despite of the placing of the antenna chip, the wireless range of the antenna is decrea-

sed. Next to interference of the radiation pattern, this can also be caused by a different

substrate (FR4) compared to the AIR Module Boosterpack [43].

A small network with 2 sensors is set up, and the visualisation is shown in figure 6.4.

Every sensor has its own colour on the graph. If a button is pressed on a sensor module

a circle will appear in the corresponding colour. The same happens when a fall occurs

(star). The last received data is available in the section ’Current data’.

6.3 Performance and measurements 93

Figure 6.4: Visualisation in Matlab of 2 sensor modules in a network

CONCLUSION 94

Chapter 7

Conclusion

This thesis was successful in achieving the goal of the assignment. A wireless transcei-

ver for personal communication powered by energy harvesting has been created.

In the previous chapter, it was proven that the system operates correctly. The system

uses solar and thermal energy to power the digital part of the system and to charge

the available storage components. The choice of the energy harvesting source used, is

based on the available and harvested power for the different types of energy. Never-

theless, other sources can also be used to power a sensor module but this module will

have less possibilities and a longer timeout period between transmissions.

There are still some small bugs need fixing. Improvements can still be made, as dis-

cussed in sections 3.1.3, 4.3.2 and 5.9.

To be usable in the suggested application of chapter 2, changes to the system have to

be made. The system has to be miniaturised (possibly using multiple layer PCBs), an

appealing package should be designed, which also integrates the EH elements. Special

measures have to be made to ensure a failure-prove system.

FIGURES AND LAYOUT PCBS 95

Appendix A

Figures and Layout PCBs

A.1 Boards of the complete sensor module

A.1.1 Solar board

Figure A.1: Solar energy harvesting board, PCB layout top layer

Figure A.2: Solar energy harvesting board, PCB layout bottom layer

A.1 Boards of the complete sensor module 96

Figure A.3: Solar energy harvesting board

A.1.2 Thermal board

Figure A.4: Thermal energy harvesting board, PCB layout top layer

Figure A.5: Thermal energy harvesting board, PCB layout bottom layer

A.1 Boards of the complete sensor module 97

Figure A.6: Thermal energy harvesting board

A.1.3 Sensor module board

Figure A.7: Sensor module board, PCB layout top layer

A.2 Test boards 98

Figure A.8: Sensor module board, PCB layout bottom layer

Figure A.9: Sensor module board

A.2 Test boards

A.2 Test boards 99

Figure A.10: Test board for solar energy harvesting (LTC3105)

Figure A.11: Test board for thermal energy harvesting (LTC3109)

A.2 Test boards 100

Figure A.12: Test board for vibrational energy harvesting (LTC3588-1)

Figure A.13: Test board for the thin-film batteries (CPC3150)

A.2 Test boards 101

Figure A.14: Active diode (LTC4413)

BIBLIOGRAPHY 102

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